Communication system

ABSTRACT

At the transmitter side, carrier waves are modulated according to an input signal for producing relevant signal points in a signal space diagram. The input signal is divided into, two, first and second, data streams. The signal points are divided into signal point groups to which data of the first data stream are assigned. Also, data of the second data stream are assigned to the signal points of each signal point group. A difference in the transmission error rate between first and second data streams is developed by shifting the signal points to other positions in the space diagram expressed at least in the polar coordinate system. At the receiver side, the first and/or second data streams can be reconstructed from a received signal. In TV broadcast service, a TV signal is divided by a transmitter into, low and high, frequency band components which are designated as a first and a second data streams respectively. Upon receiving the TV signal, a receiver can reproduce only the low frequency band component or both the low and high frequency band components, depending on its capability. Furthermore, a communication system based on an OFDM system is utilized for data transmission of a plurality of subchannels, wherein the subchannels are differentiated by changing the length of a guard time slot or a carrier wave interval of a symbol transmission time slot, or changing the transmission electric power of the carrier.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a communication system fortransmission/reception of a digital signal through modulation of itscarrier wave and demodulation of the modulated signal.

2. Description of the Prior Art

Digital signal communication systems have been used in various fields.Particularly, digital video signal transmission techniques have beenimproved remarkably.

Among them is a digital TV signal transmission method. So far, suchdigital TV signal transmission system are in particular use for e.g.transmission between TV stations. They will soon be utilized forterrestrial and/or satellite broadcast service in every country of theworld.

The TV broadcast systems including HDTV, PCM music, FAX, and otherinformation service are now demanded to increase desired data inquantity and quality for satisfying millions qf sophisticated viewers.In particular, the data has to be increased in a given bandwidth offrequency allocated for TV broadcast service. The data to be transmittedis always abundant and provided as much as handled with up-to-datetechniques of the time. It is ideal to modify or change the existingsignal transmission system corresponding to an increase in the dataamount with time.

However, the TV broadcast service is a public business and cannot gofurther without considering the interests and benefits of viewers. It isessential to have any new service appreciable with existing TV receiversand displays. More particularly, the compatibility of a system is muchdesired for providing both old and new services simultaneously or onenew service which can be intercepted by either of the existing andadvanced receivers.

It is understood that any new digital TV broadcast system to beintroduced has to be arranged for data extension in order to respond tofuture demands and technological advantages and also, for compatibleaction to allow the existing receivers to receive transmissions.

The expansion capability and compatible performance of prior art digitalTV system will be explained.

A digital satellite TV system is known in which NTSC TV signalscompressed to an about 6 Mbps are multiplexed by time divisionmodulation of 4 PSK and transmitted on 4 to 20 channels while HDTVsignals are carried on a single channel. Another digital HDTV system isprovided in which HDTV video data compressed to as small as 15 Mbps aretransmitted on a 16 or 32 QAM signal through ground stations.

Such a known satellite system permits HDTV signals to be carried on onechannel by a conventional manner, thus occupying a band of frequenciesequivalent to same channels of NTSC signals. This causes thecorresponding NTSC channels to be unavailable during transmission of theHDTV signal. Also, the compatibility between NTSC and HDTV receivers ordisplays is hardly concerned and data expansion capability needed formatching a future advanced mode is utterly disregarded.

Such a common terrestrial HDTV system offers an HDTV service onconventional 16 or 32 QAM signals without any modification. In anyanalogue TV broadcast service, there are developed a lot of signalattenuating or shadow regions within its service area due to structuralobstacles, geographical inconveniences, or signal interference from aneighbor station. When the TV signal is an analogue form, it can beintercepted more or less at such signal attenuating regions although itsreproduced picture is low in quality. If TV signal is a digital form, itcan rarely be reproduced at an acceptable level within the regions. Thisdisadvantage is critically hostile to the development of any digital TVsystem.

SUMMARY OF THE INVENTION

It is an object of the present invention, for solving the foregoingdisadvantages, to provide a communication system arranged for compatibleuse for both the existing NTSC and introducing HDTV broadcast services,particularly via satellite and also, for minimizing signal attenuatingor shadow regions of its service area on the grounds.

A communication system according to the present invention intentionallyvaries signal points, which used to be disposed at uniform intervals, toperform the signal transmission/reception. For example, if applied to aQAM-signal, the communication system comprises two major sections: atransmitter having a signal input circuit, a modulator circuit forproducing m numbers of signal points, in a signal vector field throughmodulation of a plurality of out-of-phase carrier waves using an inputsignal supplied from the input circuit, and a transmitter circuit fortransmitting a resultant modulated signal; and a receiver having aninput circuit for receiving the modulated signal, a demodulator circuitfor demodulating one-bit signal points of a QAM carrier wave, and anoutput circuit.

In operation, the input signal containing a first data stream of nvalues and a second data stream is fed to the modulator circuit of thetransmitter where a modified m-bit QAM carrier wave is producedrepresenting m signal points in a vector field. The m signal points aredivided into n signal point groups to which the n values of the firstdata stream are assigned respectively. Also, data of the second datastream are assigned to m/n signal points or sub groups of each signalpoint group. Then, a resultant transmission signal is transmitted fromthe transmitter circuit. Similarly, a third data stream can bepropagated.

At the p-bit demodulator circuit, p>m, of the receiver, the first datastream of the transmission signal is first demodulated through dividingp signal points in a signal space diagram into n signal point groups.Then, the second data stream is demodulated through assigning p/n valuesto p/n signal points of each corresponding signal point group forreconstruction of both the first and second data streams. If thereceiver is at P=n, the n signal point groups are reclaimed and assignedthe n values for demodulation and reconstruction of the first datastream.

Upon receiving the same transmission signal from the transmitter, areceiver equipped with a large sized antenna and capable of large-datamodulation can reproduce both the first and second data streams. Areceiver equipped with a small sized antenna and capable of small-datamodulation can reproduce the first data stream only. Accordingly, thecompatibility of the signal transmission system will be ensured. Whenthe first data stream is an NTSC TV signal or low frequency bandcomponent of an HDTV signal and the second data stream is a highfrequency band component of the HDTV signal, the small-data modulationreceiver can reconstruct the NTSC TV signal and the large-datamodulation receiver can reconstruct the HDTV signal. As understood, adigital NTSC/HDTV simultaneously broadcast service will be feasibleusing the compatibility of the signal transmission system of the presentinvention.

More specifically, the communication system of the present inventioncomprises: a transmitter having a signal input circuit, a modulatorcircuit for producing m signal points, in a signal vector field throughmodulation of a plurality of out-of-phase carrier waves using an inputsignal supplied from the input, and a transmitter circuit fortransmitting a resultant modulated signal, in which the main procedureincludes receiving an input signal containing a first data stream of nvalues and a second data stream, dividing the m signal points of thesignal into n signal point groups, assigning the n values of the firstdata stream to the n signal point groups respectively, assigning data ofthe second data stream to the signal points of each signal point grouprespectively, and transmitting the resultant modulated signal; and areceiver having an input circuit for receiving the modulated signal, ademodulator circuit for demodulating p signal points of a QAM carrierwave, and an output circuit, in which the main procedure includesdividing the p signal points into n signal point groups, demodulatingthe first data stream of which n values are assigned to the n signalpoint groups respectively, and demodulating the second data stream ofwhich p/n values are assigned to p/n signal points of each signal pointgroup respectively. For example, a transmitter 1 produces a modifiedm-bit QAM signal of which first, second, and third data streams, eachcarrying n values, are assigned to relevant signal point groups with amodulator 4. The signal can be intercepted and reproduced the first datastream only by a first receiver 23, both the first and second datastreams by a second receiver 33, and all the first, second, and thirdstreams by a third receiver 43.

More particularly, a receiver capable of demodulation of n-bit data canreproduce n bits from a multiple-bit modulated carrier wave carrying anm-bit data where m>n, thus allowing the communication system to havecompatibility and capability of future extension. Also, a multi-levelsignal transmission will be possible by shifting the signal points ofQAM so that a nearest signal point to the origin point of I-axis andQ-axis coordinates is spaced nf from the origin where f is the distanceof the nearest point from each axis and n is more than 1.

Accordingly, a compatible digital satellite broadcast service for boththe NTSC and HDTV systems will be feasible when the first data streamcarries an NTSC signal and the second data stream carries a differencesignal between NTSC and HDTV. Hence, the capability of corresponding toan increase in the data amount to be transmitted will be ensured. Also,at the ground, its service area will be increased while signalattenuating areas are decreased.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic view of the entire arrangement of a signaltransmission system showing a first embodiment of the present invention;

FIG. 2 is a block diagram of a transmitter of the first embodiment;

FIG. 3 is a vector diagram showing a transmission signal of the firstembodiment;

FIG. 4 is a vector diagram showing a transmission signal of the firstembodiment;

FIG. 5 is a view showing an assignment of binary codes to signal pointsaccording to the first embodiment;

FIG. 6 is a view showing an assignment of binary codes to signal pointgroups according to the first embodiment;

FIG. 7 is a view showing an assignment of binary codes to signal pointsin each signal point group according to the first embodiment;

FIG. 8 is a view showing another assignment of binary codes to signalpoint groups and their signal points according to the first embodiment;

FIG. 9 is a view showing threshold values of the signal point groupsaccording to the first embodiment;

FIG. 10 is a vector diagram of a modified 16 QAM signal of the firstembodiment;

FIG. 11 is a graphic diagram showing the relation between antenna radiusr₂ and transmission energy ratio n according to the first embodiment;

FIG. 12 is a view showing the signal points of a modified 64 QAM signalof the first embodiment;

FIG. 13 is a graphic diagram showing the relation between antenna radiusr₃ and transmission energy ratio n according to the first embodiment;

FIG. 14 is a vector diagram showing signal point groups and their signalpoints of the modified 64 QAM signal of the first embodiment;

FIG. 15 is an explanatory view showing the relation between A₁ and A₂ ofthe modified 64 QAM signal of the first embodiment;

FIG. 16 is a graph diagram showing the relation between antenna radiusr₂, r₃ and transmission energy ratio n₁₆, n₆₄ respectively according tothe first embodiment;

FIG. 17 is a block diagram of a digital transmitter of the firstembodiment;

FIG. 18 is a signal space diagram of a 4 PSK modulated signal of thefirst embodiment;

FIG. 19 is a block diagram of a first receiver of the first embodiment;

FIG. 20 is a signal space diagram of a 4 PSK modulated signal of thefirst embodiment;

FIG. 21 is a block diagram of a second receiver of the first embodiment;

FIG. 22 is a vector diagram of a modified 16 QAM signal of the firstembodiment;

FIG. 23 is a vector diagram of a modified 64 QAM signal of the firstembodiment;

FIG. 24 is a flow chart showing an action of the first embodiment;

FIGS. 25(a) and 25(b) are vector diagrams showing an 8 and a 16 QAMsignal of the first embodiment respectively;

FIG. 26 is a block diagram of a third receiver of the first embodiment;

FIG. 27 is a view showing signal points of the modified 64 QAM signal ofthe first embodiment;

FIG. 28 is a flow chart showing another action of the first embodiment;

FIG. 29 is a schematic view of the entire arrangement of a signaltransmission system showing a third embodiment of the present invention;

FIG. 30 is a block diagram of a first video encoder of the thirdembodiment;

FIG. 31 is a block diagram of a first video decoder of the thirdembodiment;

FIG. 32 is a block diagram of a second video decoder of the thirdembodiment;

FIG. 33 is a block diagram of a third video decoder of the thirdembodiment;

FIG. 34 is an explanatory view showing a time multiplexing of D₁, D₂,and D₃ signals according to the third embodiment;

FIG. 35 is an explanatory view showing another time multiplexing of theD₁, D₂, and D₃ signals according to the third embodiment;

FIG. 36 is an explanatory view showing a further time multiplexing ofthe D₁, D₂, and D₃ signals according to the third embodiment;

FIG. 37 is a schematic view of the entire arrangement of a signaltransmission system showing a fourth embodiment of the presentinvention;

FIG. 38 is a vector diagram of a modified 16 QAM signal of the thirdembodiment;

FIG. 39 is a vector diagram of the modified 16 QAM signal of the thirdembodiment;

FIG. 40 is a vector diagram of a modified 64 QAM signal of the thirdembodiment;

FIG. 41 is a diagram of assignment of data components on a time baseaccording to the third embodiment;

FIG. 42 is a diagram of assignment of data components on a time base inTDMA action according to the third embodiment;

FIG. 43 is a block diagram of a carrier reproducing circuit of the thirdembodiment;

FIG. 44 is a diagram showing the principle of carrier wave reproductionaccording to the third embodiment;

FIG. 45 is a block diagram of a carrier reproducing circuit for reversemodulation of the third embodiment;

FIG. 46 is a diagram showing an assignment of signal points of the 16QAM signal of the third embodiment;

FIG. 47 is a diagram showing an assignment of signal points of the 64QAM signal of the third embodiment;

FIG. 48 is a block diagram of a carrier reproducing circuit for 16×multiplication of the third embodiment;

FIG. 49 is an explanatory view showing a time multiplexing of D_(V1),D_(H1), D_(V2), D_(H2) , D_(V3), and D_(H3) signals according to thethird embodiment;

FIG. 50 is an explanatory view showing a TDMA time multiplexing ofD_(V1), D_(H1), D_(V2), D_(H2), D_(V3), and D_(H3) signals according tothe third embodiment;

FIG. 51 is an explanatory view showing another TDMA time multiplexing ofthe D_(V1), D_(H1), D_(V2), D_(H2), D_(V3), and D_(H3) signals accordingto the third embodiment;

FIG. 52 is a diagram showing a signal interference region in a knowntransmission method according to the fourth embodiment;

FIG. 53 is a diagram showing signal interference regions in amulti-level signal transmission method according to the fourthembodiment;

FIG. 54 is a diagram showing signal attenuating regions in the knowntransmission method according to the fourth embodiment;

FIG. 55 is a diagram showing signal attenuating regions in themulti-level signal transmission method according to the fourthembodiment;

FIG. 56 is a diagram showing a signal interference region between twodigital TV stations according to the fourth embodiment;

FIG. 57 is a diagram showing an assignment of signal points of amodified 4 ASK signal of the fifth embodiment;

FIG. 58 is a diagram showing another assignment of signal points of themodified 4 ASK signal of the fifth embodiment;

FIGS. 59(a) and 59(b) are diagrams showing assignment of signal pointsof the modified 4 ASK signal of the fifth embodiment;

FIG. 60 is a diagram showing another assignment of signal points of themodified 4 ASK signal of the fifth embodiment when the C/N rate is low;

FIG. 61 is a block diagram of a transmitter of the fifth embodiment;

FIGS. 62(a) and 62(b) are diagrams showing frequency distributionprofiles of an ASK modulated signal of the fifth embodiment;

FIG. 63 is a block diagram of a receiver of the fifth embodiment;

FIG. 64 is a block diagram of a video signal transmitter of the fifthembodiment;

FIG. 65 is a block diagram of a TV receiver of the fifth embodiment;

FIG. 66 is a block diagram of another TV receiver of the fifthembodiment;

FIG. 67 is a block diagram of a satellite-to-ground TV receiver of thefifth embodiment;

FIG. 68 is a diagram showing an assignment of signal points of an 8 ASKsignal of the fifth embodiment;

FIG. 69 is a block diagram of a video encoder of the fifth embodiment;

FIG. 70 is a block diagram of a video encoder of the fifth embodimentcontaining one divider circuit;

FIG. 71 is a block diagram of a video decoder of the fifth embodiment;

FIG. 72 is a block diagram of a video decoder of the fifth embodimentcontaining one mixer circuit;

FIG. 73 is a diagram showing a time assignment of data components of atransmission signal according to the fifth embodiment;

FIG. 74(a) is a block diagram of a video decoder of the fifthembodiment;

FIG. 74(b) is a diagram showing another time assignment of datacomponents of the transmission signal according to the fifth embodiment;

FIG. 75 is a diagram showing a time assignment of data components of atransmission signal according to the fifth embodiment;

FIG. 76 is a diagram showing a time assignment of data components of atransmission signal according to the fifth embodiment;

FIG. 77 is a diagram showing a time assignment of data components of atransmission signal according to the fifth embodiment;

FIG. 78 is a block diagram of a video decoder of the fifth embodiment;

FIG. 79 is a diagram showing a time assignment of data components of athree-level transmission signal according to the fifth embodiment;

FIG. 80 is a block diagram of another video decoder of the fifthembodiment;

FIG. 81 is a diagram showing a time assignment of data components of atransmission signal according to the fifth embodiment;

FIG. 82 is a block diagram of a video decoder for D₁ signal of the fifthembodiment;

FIG. 83 is a graphic diagram showing the relation between frequency andtime of a frequency modulated signal according to the fifth embodiment;

FIG. 84 is a block diagram of a magnetic record/playback apparatus ofthe fifth embodiment;

FIG. 85 is a graphic diagram showing the relation between C/N and levelaccording to the second embodiment;

FIG. 86 is a graphic diagram showing the relation between C/N andtransmission distance according to the second embodiment;

FIG. 87 is a block diagram of a transmission of the second embodiment;

FIG. 88 is a block diagram of a receiver of the second embodiment;

FIG. 89 is a graphic diagram showing the relation between C/N and errorrate according to the second embodiment;

FIG. 90 is a diagram showing signal attenuating regions in thethree-level transmission of the fifth embodiment;

FIG. 91 is a diagram showing signal attenuating regions in thefour-level transmission of a sixth embodiment;

FIG. 92 is a diagram showing the four-level transmission of the sixthembodiment;

FIG. 93 is a block diagram of a divider of the sixth embodiment;

FIG. 94 is a block diagram of a mixer of the sixth embodiment;

FIG. 95 is a diagram showing another four-level transmission of thesixth embodiment;

FIG. 96 is a view of signal propagation of a known digital TV broadcastsystem;

FIG. 97 is a view of signal propagation of a digital TV broadcast systemaccording to the sixth embodiment;

FIG. 98 is a diagram showing a four-level transmission of the sixthembodiment;

FIG. 99 is a vector diagram of a 16 SRQAM signal of the thirdembodiment;

FIG. 100 is a vector diagram of a 32 SRQAM signal of the thirdembodiment;

FIG. 101 is a graphic diagram showing the relation between C/N and errorrate according to the third embodiment;

FIG. 102 is a graphic diagram showing the relation between C/N and errorrate according to the third embodiment;

FIG. 103 is a graphic diagram showing the relation between shiftdistance n and C/N needed for transmission according to the thirdembodiment;

FIG. 104 is a graphic diagram showing the relation between shiftdistance n and C/N needed for transmission according to the thirdembodiment;

FIG. 105 is a graphic diagram showing the relation between signal leveland distance from a transmitter antenna in terrestrial broadcast serviceaccording to the third embodiment;

FIG. 106 is a diagram showing a service area of the 32 SRQAM signal ofthe third embodiment;

FIG. 107 is a diagram showing a service area of the 32 SRQAM signal ofthe third embodiment;

FIG. 108(a) is a diagram showing a frequency distribution profile of aconventional TV signal, FIG. 108(b) is a diagram showing a frequencydistribution profile of a conventional two-layer TV signal, FIG. 108(c)is a diagram showing threshold values of the third embodiment, FIG.108(d) is a diagram showing a frequency distribution profile oftwo-layer OFDM carriers of the ninth embodiment, and FIG. 108(e) is adiagram showing threshold values for three-layer OFDM of the ninthembodiment;

FIG. 109 is a diagram showing a time assignment of the TV signal of thethird embodiment;

FIG. 110 is a diagram showing a principle of C-CDM of the thirdembodiment;

FIG. 111 is a view showing an assignment of codes according to the thirdembodiment;

FIG. 112 is a view showing an assignment of an extended 36 QAM accordingto the third embodiment;

FIG. 113 is a view showing a frequency assignment of a modulation signalaccording to the fifth embodiment;

FIG. 114 is a block diagram showing a magnetic recording/playbackapparatus according to the fifth embodiment;

FIG. 115 is a block diagram showing a transmitter/receiver of a portabletelephone according to the eighth embodiment;

FIG. 116 is a block diagram showing base stations according to theeighth embodiment;

FIG. 117 is a view illustrating communication capacities and trafficdistribution of a conventional system;

FIG. 118 is a view illustrating communication capacities and trafficdistribution according to the eighth embodiment;

FIG. 119(a) is a diagram showing a time slot assignment of aconventional system;

FIG. 119(b) is a diagram showing a time slot assignment according to theeighth embodiment;

FIG. 120(a) is a diagram showing a time slot assignment of aconventional TDMA system;

FIG. 120(b) is a diagram showing a time slot assignment according to aTDMA system of the eighth embodiment;

FIG. 121 is a block diagram showing a one-level transmitter/receiveraccording to the eighth embodiment;

FIG. 122 is a block diagram showing a two-level transmitter/receiveraccording to the eighth embodiment;

FIG. 123 is a block diagram showing an OFDM type transmitter/receiveraccording to the ninth embodiment;

FIG. 124 is a view illustrating a principle of the OFDM system accordingto the ninth embodiment;

FIG. 125(a) is a view showing a frequency assignment of a modulationsignal of a conventional system;

FIG. 125(b) is a view showing a frequency assignment of a modulationsignal according to the ninth embodiment;

FIG. 126(a) is a view showing a frequency assignment of an OFDM signalof the ninth embodiment, wherein no weighting is applied;

FIG. 126(b) is a view showing a frequency assignment of an OFDM signalof the ninth embodiment, wherein two channels of two-layer OFDM areweighted by transmission electric power;

FIG. 126(c) is a view showing a frequency assignment of an OFDM signalof the ninth embodiment, wherein carrier intervals are doubled byweighting;

FIG. 126(d) is a view showing a frequency assignment of an OFDM signalof the ninth embodiment, wherein carrier intervals are not weighted;

FIG. 127 is a block diagram showing a transmitter/receiver according tothe ninth embodiment;

FIG. 128 is a block diagram showing a Trellis encoder according to thefifth embodiment;

FIG. 129 is a view showing a time assignment of effective symbol periodsand guard intervals according to the ninth embodiment;

FIG. 130 is a graphic diagram showing a relation between C/N rate anderror rate according to the ninth embodiment;

FIG. 131 is a block diagram showing a magnetic recording/playbackapparatus according to the fifth embodiment;

FIG. 132 is a view showing a recording format of track on the magnetictape and a travelling of a head;

FIG. 133 is a block diagram showing a transmitter/receiver according tothe third embodiment;

FIG. 134 is a diagram showing a frequency assignment of a conventionalbroadcasting;

FIG. 135 is a diagram showing a relation between service area andpicture quality in a three-level signal transmission system according tothe third embodiment;

FIG. 136 is a diagram showing a frequency assignment in case themulti-level signal transmission system according to the third embodimentis combined with FDM;

FIG. 137 is a block diagram showing a transmitter/receiver according tothe third embodiment, in which Trellis encoding is adopted;

FIG. 138 is a block diagram showing a transmitter/receiver according tothe ninth embodiment, in which a part of low frequency band signal istransmitted by OFDM;

FIG. 139 is a diagram showing an assignment of signal points of the8-PS-APSK signal of the first embodiment;

FIG. 140 is a diagram showing an assignment of signal points of the16-PS-APSK signal of the first embodiment;

FIG. 141 is a diagram showing an assignment of signal points of the8-PS-PSK signal of the first embodiment;

FIG. 142 is a diagram showing an assignment of signal points of the16-PS-PSK (PS type) signal of the first embodiment;

FIG. 143 is a graphic diagram showing the relation between antennaradius of satellite and transmission capacity according to the firstembodiment;

FIG. 144 is a block diagram showing a weighted OFDM transmitter/receiveraccording to the ninth embodiment;

FIG. 145(a) is a diagram showing the waveform of the guard time and thesymbol time in the multi-level OFDM according to the ninth embodiment,wherein multipath is short;

FIG. 145(b) is a diagram showing the waveform of the guard time and thesymbol time in the multi-level OFDM according to the ninth embodiment,wherein multipath is long;

FIG. 146 is a diagram showing a principle of the multi-level OFDMaccording to the ninth embodiment;

FIG. 147 is a diagram showing subchannel assignment of a two-layersignal transmission system, weighted by electric power according to theninth embodiment;

FIG. 148 is a diagram showing relation among the D/V ratio, themultipath delay time, and the guard time according to the ninthembodiment;

FIG. 149(a) is a diagram showing time slots of respective layersaccording to the ninth embodiment;

FIG. 149(b) is a diagram showing time distribution of guard times ofrespective layers according to the ninth embodiment;

FIG. 149(c) is a diagram showing time distribution of guard times ofrespective layers according to the ninth embodiment;

FIG. 150 is a diagram showing relation between multipath delay time andtransfer rate according to the ninth embodiment, wherein three-layersignal transmission effective to multipath is realized; and

FIG. 151 is a diagram showing relation between multipath delay time andC/N ratio according to the ninth embodiment, wherein two-dimensional,matrix type, multi-layer broadcast service can be realized by combiningthe GTW-OFDM and the C-CDM (or the CSW-OFDM).

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS Embodiment 1

One embodiment of the present invention will be described referring tothe relevant drawings.

FIG. 1 shows the entire arrangement of a signal transmission systemaccording to the present invention. A transmitter 1 comprises an inputunit 2, a divider circuit 3, a modulator 4, and a transmitter unit 5. Inaction, each input multiplex signal is divided by the divider circuit 3into three groups, a first data stream D1, a second data stream D2, athird data stream D3, which are then modulated by the modulator 4 beforetransmitted from the transmitter unit 5. The modulated signal is sent upfrom an antenna 16 through an uplink 7 to a satellite 10 where it isintercepted by an uplink antenna 11 and amplified by a transponder 12before transmitted from a downlink antenna 13 towards the ground.

The transmission signal is then sent down through three downlinks 21,32, and 41 to a first 23, a second 33, and a third receiver 43respectively. In the first receiver 23, the signal intercepted by anantenna 22 is fed through an input unit 24 to a demodulator 25 where itsfirst data stream only is demodulated, while the second and third datastreams are not recovered, before transmitted further from an outputunit 26.

Similarly, the second receiver 33 allows the first and second datastreams of the signal intercepted by an antenna 32 and fed from an inputunit 34 to be demodulated by a demodulator 35 and then, summed by asummer 37 to a single data stream which is then transmitted further froman output unit 36.

The third receiver 43 allows all the first, second, and third datastreams of the signal intercepted by an antenna 42 and fed from an inputunit 44 to be demodulated by a demodulator 45 and then, summed by asummer 47 to a single data stream which is then transmitted further froman output unit 46.

As understood, the three discrete receivers 23, 33, and 43 have theirrespective demodulators of different characteristics such that theiroutputs demodulated from the same frequency band signal of thetransmitter 1 contain data of different sizes. More particularly, threedifferent but compatible data can simultaneously be carried on a givenfrequency band signal to their respective receivers. For example, eachof three, existing NTSC, HDTV, and super HDTV, digital signals isdivided into a low, a high, and a super high frequency band componentswhich represent the first, the second, and the third data streamrespectively. Accordingly, the three different TV signals can betransmitted on a one-channel frequency band carrier for simultaneousreproduction of a medium, a high, and a super high resolution TV imagerespectively.

In service, the NTSC TV signal is intercepted by a receiver accompaniedwith a small antenna for demodulation of a small-sized data, the HDTVsignal is intercepted by a receiver accompanied with a medium antennafor demodulation of medium-sized data, and the super HDTV signal isintercepted by a receiver accompanied with a large antenna fordemodulation of large-sized data. Also, as illustrated in FIG. 1, adigital NTSC TV signal containing only the first data stream for digitalNTSC TV broadcasting service is fed to a digital transmitter 51 where itis received by an input unit 52 and modulated by a demodulator 54 beforetransmitted further from a transmitter unit 55. The demodulated signalis then sent up from an antennal 56 through an uplink 57 to thesatellite 10 which in turn transmits the same through a downlink 58 tothe first receiver 23 on the ground.

The first receiver 23 demodulates with its demodulator 25 the modulateddigital signal supplied from the digital transmitter 51 to the originalfirst data stream signal. Similarly, the same modulated digital signalcan be intercepted and demodulated by the second 33 or third receiver 42to the first data stream or NTSC TV signal. In summary, the threediscrete receivers 23, 33, and 43 all can intercept and process adigital signal of the existing TV system for reproduction.

The arrangement of the signal transmission system will be described inmore detail.

FIG. 2 is a block diagram of the transmitter 1, in which an input signalis fed across the input unit 2 and divided by the divider circuit 3 intothree digital signals containing a first, a second, and a third datastream respectively.

Assuming that the input signal is a video signal, its low frequency bandcomponent is assigned to the first data stream, its high frequency bandcomponent to the second data stream, its super-high frequency bandcomponent to the third data stream. The three different frequency bandsignals are fed to a modulator input 61 of the modulator 4. Here, asignal point modulating/changing circuit 67 modulates or changes thepositions of the signal points according to an externally given signal.The modulator 4 is arranged for amplitude modulation on two90°-out-of-phase carriers respectively which are then summed to amultiple QAM signal. More specifically, the signal from the modulatorinput 61 is fed to both a first 62 and a second AM modulator 63. Also, acarrier wave of cos(2πfct) produced by a carrier generator 64 isdirectly fed to the first AM modulator 62 and also, to a π/2 phaseshifter 66 where it is 90° shifted in phase to a sin(2πfct) form priorto transmitted to the second AM modulator 63. The two amplitudemodulated signals from the first and second AM modulators 62, 63 aresummed by a summer 65 to a transmission signal which is then transferredto the transmitter unit 5 for output. The procedure is well known andwill no further be explained.

The QAM signal will now be described in a common 8×8 or 16 stateconstellation referring to the first quadrant of a space diagram in FIG.3. The output signal of the modulator 4 is expressed by a sum vector oftwo, A cos 2πfct and B cos 2πfct, vectors 81, 82 which represent the two90°-out-of-phase carriers respectively. When the distal point of a sumvector from the zero point represents a signal point, the 16 QAM signalhas 16 signal points determined by a combination of four horizontalamplitude values a₁, a₂, a₃, a₄ and four vertical amplitude values b₁,b₂, b₃, b₄. The first quadrant in FIG. 3 contains four signal points 83at C₁₁, 84 at C₁₂, 85 at C₂₂, and 86 at C₂₁.

C₁₁ is a sum vector of a vector 0-a₁ and a vector 0-b₁ and thus,expressed as C₁ =a₁ cos 2πfct-b sin 2πfct=A cos(2πfct+dπ/2).

It is now assumes that the distance between 0 and a₁ in the orthogonalcoordinates of FIG. 3 is A₁, between a₁ and a₂ is A₂, between 0 and b₁is B₁, and between b₁ and b₂ is B₂.

As shown in FIG. 4, the 16 signal points are allocated in a vectorcoordinate, in which each point represents a four-bit pattern thus toallow the transmission of four bit data per period or time slot.

FIG. 5 illustrates a common assignment of two-bit patterns to the 16signal points.

When the distance between two adjacent signal points is great, it willbe identified by the receiver with much ease. Hence, it is desired tospace the signal points at greater intervals. If two particular signalpoints are allocated near to each other, they are rarely distinguishedand error rate will be increased. Therefore, it is most preferred tohave the signal points spaced at equal intervals as shown in FIG. 5, inwhich the 16 QAM signal is defined by A₁=A₂/2.

The transmitter 1 of the embodiment is arranged to divide an inputdigital signal into a first, a second, and a third data or bit stream.The 16 signal points or groups of signal points are divided into fourgroups. Then, 4 two-bit patterns of the first data stream are assignedto the four signal point groups respectively, as shown in FIG. 6. Moreparticularly, when the two-bit pattern of the first data stream is 11,one of four signal points of the first signal point group 91 in thefirst quadrant is selected depending on the content of the second datastream for transmission. Similarly, when 01, one signal point of thesecond signal point group 92 in the second quadrant is selected andtransmitted. When 00, one signal point of the third signal point group93 in the third quadrant is transmitted and when 10, one signal point ofthe fourth signal point group 94 in the fourth quadrant is transmitted.Also, 4 two-bit patterns in the second data stream of the 16 QAM signal,or e.g. 16 four-bit patterns in the second data stream of a 64-state QAMsignal, are assigned to four signal points or sub signal point groups ofeach of the four signal point groups 91, 92, 93, 94 respectively, asshown in FIG. 7. It should be understood that the assignment issymmetrical between any two quadrants. The assignment of the signalpoints to the four groups 91, 92, 93, 94 is determined by priority tothe two-bit data of the first data stream. As the result, two-bit dataof the first data stream and two-bit data of the second data stream canbe transmitted independently. Also, the first data stream will bedemodulated with the use of a common 4 PSK receiver having a givenantenna sensitivity. If the antenna sensitivity is higher, a modifiedtype of the 16 QAM receiver of the present invention will intercept anddemodulate both the first and second data stream with equal success.

FIG. 8 shows an example of the assignment of the first and second datastreams in two-bit patterns.

When the low frequency band component of an HDTV video signal isassigned to the first data stream and the high frequency component tothe second data stream, the 4 PSK receiver can produce an NTSC-levelpicture from the first data stream and the 16- or 64-state QAM receivercan produce an HDTV picture from a composite reproduction signal of thefirst and second data streams.

Since the signal points are allocated at equal intervals, there isdeveloped in the 4 PSK receiver a threshold distance between thecoordinate axes and the shaded area of the first quadrant, as shown inFIG. 9. If the threshold distance is A_(T0), a PSK signal having anamplitude of A_(T0) will successfully be intercepted. However, theamplitude has to be increased to a three times greater value or 3A_(T0)for transmission of a 16 QAM signal while the threshold distance A _(TO)being maintained. More particularly, the energy for transmitting the 16QAM signal is needed nine times greater than that for sending the 4 PSKsignal. Also, when the 4 PSK signal is transmitted in a 16 QAM mode,energy waste will be high and reproduction of a carrier signal will betroublesome. Above all, the energy available for satellite transmittingis not abundant but strictly limited to minimum use. Hence, nolarge-energy-consuming signal transmitting system will be put intopractice until more energy for satellite transmission is available. Itis expected that a great number of the 4 PSK receivers are introducedinto the market as digital TV broadcasting is soon in service. Afterintroduction to the market, the 4 PSK receivers will hardly be shiftedto higher sensitivity models because a signal interceptingcharacteristic gap between the two, old and new, models is high.Therefore, the transmission of the 4 PSK signals must not be abandoned.

In this respect, a new system is desperately needed for transmitting thesignal point data of a quasi 4 PSK signal in the 16 QAM mode with theuse of less energy. Otherwise, the limited energy at a satellite stationwill degrade the entire transmission system.

The present invention resides in a multiple signal level arrangement inwhich the four signal point groups 91, 92, 93 94 are allocated at agreater distance from each other, as shown in FIG. 10, for minimizingthe energy consumption required for 16 QAM modulation of quasi 4 PSKsignals.

For clearing the relation between the signal receiving sensitivity andthe transmitting energy, the arrangement of the digital transmitter 51and the first receiver 23 will be described in more detail referring toFIG. 1. Both the digital transmitter 51 and the first receiver 23 areformed of known types for data transmission or video signal transmissione.g. in TV broadcasting service. As shown in FIG. 17, the digitaltransmitter 51 is a 4 PSK transmitter equivalent to the multiple-bit QAMtransmitter 1, shown in FIG. 2, without AM modulation capability. Inoperation, an input signal is fed through an input unit 52 to amodulator 54 where it is divided by a modulator input 121 to twocomponents. The two components are then transferred to a first two-phasemodulator circuit 122 for phase modulation of a base carrier and asecond two-phase modulator circuit 123 for phase modulation of a carrierwhich is 90° out of phase with the base carrier respectively. Twooutputs of the first and second two-phase modulator circuits 122, 123are then summed by a summer 65 to a composite modulated signal which isfurther transmitted from a transmitter unit 55.

The resultant modulated signal is shown in the space diagram of FIG. 18.

It is known that the four signal points are allocated at equal distancesfor achieving optimum energy utilization. FIG. 18 illustrates an examplewhere the four signal points 125, 126, 127, 128 represent 4 two-bitpatterns, 11, 01, 00, and 10 respectively. It is also desired forsuccessful data transfer from the digital transmitter 51 to the firstreceiver 23 than the 4 PSK signal from the digital transmitter 51 has anamplitude of not less than a given level. More specifically, when theminimum amplitude of the 4 PSK signal needed for transmission from thedigital transmitter 51 to the first receiver 23 of 4 PSK mode, or thedistance between 0 and a₁ in FIG. 18 is A_(T0), the first receiver 23successfully intercept any 4 PSK signal having an amplitude of more thanA_(T0).

The first receiver 23 is arranged to receive at its small-diameterantenna 22 a desired or 4 PSK signal which is transmitted from thetransmitter 1 or digital transmitter 51 respectively through thetransponder 12 of the satellite 10 and demodulate it with thedemodulator 24. In more particular, the first receiver 23 issubstantially designed for interception of a digital TV or datacommunications signal of 4 PSK or 2 PSK mode.

FIG. 19 is a block diagram of the first receiver 23 in which an inputsignal received by the antenna 22 from the satellite 12 is fed throughthe input unit 24 to a carrier reproducing circuit 131 where a carrierwave is demodulated and to a π/2 phase shifter 132 where a 90° phasecarrier wave is demodulated. Also, two 90°-out-of-phase components ofthe input signal are detected by a first 133 and a second phase detectorcircuit 134 respectively and transferred to a first 136 and a seconddiscrimination/demodulation circuit 137 respectively. Two demodulatedcomponents from their respective discrimination/demodulation circuits136 and 137, which have separately been discriminated at units of timeslot by means of timing signals from a timing wave extracting circuit135, are fed to a first data stream reproducing unit 232 where they aresummed to a first data stream signal which is then delivered as anoutput from the output unit 26.

The input signal to the first receiver 23 will now be explained in moredetail referring to the vector diagram of FIG. 20. The 4 PSK signalreceived by the first receiver 23 from the digital transmitter 51 isexpressed in an ideal form without transmission distortion and noise,using four signal points 151, 152, 153, 154 shown in FIG. 20.

In practice, the real four signal points appear in particular extendedareas about the ideal signal positions 151, 152, 153, 154 respectivelydue to noise, amplitude distortion, and phase error developed duringtransmission. If one signal point is unfavorably displaced from itsoriginal position, it will hardly be distinguished from its neighborsignal point and the error rate will thus be increased. As the errorrate increases to a critical level, the reproduction of data becomesless accurate. For enabling the data reproduction at a maximumacceptable level of the error rate, the distance between any two signalpoints should be far enough to be distinguished from each other. If thedistance is 1A_(R0), the signal point 151 of a 4 PSK signal at close toa critical error level has to stay in a first discriminating area 155denoted by the hatching of FIG. 20 and determined by |0−a_(R1)|≧A_(R0)and |0−_(R1)|≧A_(R0). This allows the signal transmission system toreproduce carrier waves and thus, demodulate a wanted signal. When theminimum radius of the antenna 22 is set to r₀, the transmission signalof more than a given level can be intercepted by any receiver of thesystem. The amplitude of a 4 PSK signal of the digital transmitter 51shown in FIG. 18 is minimum at A_(T0) and thus, the minimum amplitudeA_(R0) of a 4 PSK signal to be received by the first receiver 23 isdetermined equal to A_(T0). As the result, the first receiver 23 canintercept and demodulate the 4 PSK signal from the digital transmitter51 at the maximum acceptable level of the error rate when the radius ofthe antenna 22 is more than r₀. If the transmission signal is ofmodified 16- or 64-state QAM mode, the first receiver 23 may finddifficult to reproduce its carrier wave. For compensation, the signalpoints are increased to eight which are allocated at angles of(π/4+nπ/2) as shown in FIG. 25(a) and its carrier wave will bereproduced by a 16× multiplication technique. Also, if the signal pointsare assigned to 16 locations at angles of nπ/8 as shown in FIG. 25(b),the carrier of a quasi 4 PSK mode 16 QAM modulated signal can bereproduced with the carrier reproducing circuit 131 which is modifiedfor performing 16× frequency multiplication. At the time, the signalpoints in the transmitter 1 should be arranged to satisfyA₁/(A₁+A₂)=tan(π/8).

Here, a case of receiving a QPSK signal will be considered. Similarly tothe manner performed by the signal point modulating/changing circuit 67in the transmitter shown in FIG. 2, it is also possible to modulate thepositions of the signal points of the QPSK signal shown in FIG. 18(amplitude-modulation, pulse-modulation, or the like). In this case, thesignal point demodulating unit 138 in the first receiver 23 demodulatesthe position modulated or position changed signal. The demodulatedsignal is outputted together with the first data stream.

The 16 PSK signal of the transmitter 1 will now be explained referringto the vector diagram of FIG. 9. When the horizontal vector distance A₁of the signal point 83 is greater than A_(T0) of the minimum amplitudeof the 4 PSK signal of the digital transmitter 51, the four signalpoints 83, 84, 85, 86 in the first quadrant of FIG. 9 stay in the shadedor first 4 PSK signal receivable area 87. When received by the firstreceiver 23, the four points of the signal appear in the firstdiscriminating area of the vector field shown in FIG. 20. Hence, any ofthe signal points 83, 84, 85, 86 of FIG. 9 can be translated into thesignal level 151 of FIG. 20 by the first receiver 23 so that the two-bitpattern of 11 is assigned to a corresponding time slot. The two-bitpattern of 11 is identical to 11 of the first signal point group 91 orfirst data stream of a signal from the transmitter 1. Equally, the firstdata stream will be reproduced at the second, third, or fourth quadrant.As the result, the first receiver 23 reproduces two-bit data of thefirst data stream out of the plurality of data streams in a 16-, 32-, or64-state QAM signal transmitted from the transmitter 1. The second andthird data streams are contained in four segments of the signal pointgroup 91 and thus, will not affect on the demodulation of the first datastream. They may however affect the reproduction of a carrier wave andan adjustment, described later, will be needed.

If the transponder of a satellite supplies an abundance of energy, theforgoing technique of 16 to 64-state QAM mode transmission will befeasible. However, the transponder of the satellite in any existingsatellite transmission system is strictly limited in the power supplydue to its compact size and the capability of solar batteries. If thetransponder or satellite is increased in size thus weight, its launchingcost will soar. This disadvantage will rarely be eliminated bytraditional techniques unless the cost of launching a satellite rocketis reduced to a considerable level. In the existing system, a commoncommunications satellite provides as low as 20 W of power supply and acommon broadcast satellite offers 100 W to 200 W at best. Fortransmission of such a 4 PSK signal in the symmetrical 16-state QAM modeas shown in FIG. 9, the minimum signal point distance is needed 3A_(T0)as the 16 QAM amplitude is expressed by 2A₁=A₂. Thus, the energy neededfor the purpose is nine times greater than that for transmission of acommon 4 PSK signal, in order to maintain compatibility. Also, anyconventional satellite transponder can hardly provide a power forenabling such a small antenna of the 4 PSK first receiver to intercept atransmitted signal therefrom. For example, in the existing 40 W system,360 W is needed for appropriate signal transmission and will beunrealistic in the respect of cost.

It would be under stood that the symmetrical signal state QAM techniqueis most effective when the receivers equipped with the same sizedantennas are employed corresponding to a given transmitting power.Another novel technique will however be preferred for use with thereceivers equipped with different sized antennas.

In more detail, while the 4 PSK signal can be intercepted by a commonlow cost receiver system having a small antenna, the 16 QAM signal isintended to be received by a high cost, high quality, multiple-bitmodulating receiver system with a medium or large sized antenna which isdesigned for providing highly valuable services, e.g. HDTVentertainments, to a particular person who invests more money. Thisallows both 4 PSK and 16 QAM signals, if desired, with a 64 DMA, to betransmitted simultaneously with the help of a small increase in thetransmitting power.

For example, the transmitting power can be maintained low when thesignal points are allocated at A₁=A₂ as shown in FIG. 10. The amplitudeA(4) for transmission of 4 PSK data is expressed by a vector 96equivalent to a square root of (A₁+A₂)²+(B₁+B₂)². Then,|A(4)|² =A ₁ ² +B ₁ ² =A _(T0) ² +A _(T0) ²=2A _(T0) ²|A(16)|²=(A ₁ +A ₂)²+(B ₁ +B ₂)²=4A _(T0) ²+4A _(T0) ²=8_(T0) ²|A(16)|/|A(4)|=2

Accordingly, the 16 QAM signal can be transmitted at a two times greateramplitude and a four times greater-transmitting energy than those neededfor the 4 PSK signal. A modified 16 QAM signal according to the presentinvention will not be demodulated by a common receiver designed forsymmetrical, equally distanced signal point QAM. However, it can bedemodulated with the second receiver 33 when two threshold A₁ and A₂ arepredetermined to appropriate values. At FIG. 10, the minimum distancebetween two signal points in the first segment of the signal point group91 is A₁ and A₂/2A₁ is established as compared with the distance 2A₁ of4 PSK. Then, as A₁=A₂, the distance becomes ½. This explains that thesignal receiving sensitivity has to be two times greater for the sameerror rate and four times greater for the same signal level. For havinga four times greater value of sensitivity, the radius r₂ of the antenna32 of the second receiver 33 has to be two times greater than the radiusr₁ of the antenna 22 of the first receiver 23 thus satisfying r₂'2 r ₁.For example, the antenna 32 of the second receiver 33 is 60 cm diameterwhen the antenna 22 if the first receiver 23 is 30 cm. In this manner,the second data stream representing the high frequency component of anHDTV will be carried on a signal channel and demodulated successfully.As the second receiver 33 intercepts the second data stream or a higherdata signal, its owner can enjoy a return of high investment. Hence, thesecond receiver 33 of a high price may be accepted. As the minimumenergy for transmission of 4 PSK data is predetermined, the ratio n₁₆ ofmodified 16 APSK transmitting energy to 4 PSK transmitting energy willbe calculated to the antenna radius r₂ of the second receiver 33 using aratio between A₁ and A₂ shown in FIG. 10.

In particular, n₁₆ is expressed by ((A₁+A₂)/A₁)² which is the minimumenergy for transmission of 4 PSK data. As the signal point distancesuited for modified 16 QAM interception is A₂, the signal point distancefor 4 PSK interception is 2A₁, and the signal point distance ratio isA₂/2A₁, the antenna radius r₂ is determined as shown in FIG. 11, inwhich the curve 101 represents the relation between the transmittingenergy ratio n₁₆ and the radius r₂ of the antenna 22 of the secondreceiver 23.

Also, the point 102 indicates transmission of common 16 QAM at the equaldistance signal state mode where the transmitting energy is nine timesgreater and thus will no more be practical. As apparent from the graphof FIG. 11, the antenna radius r₂ of the second receiver 23 cannot bereduced further even if n₁₆ is increased more than 5 times.

The transmitting energy at the satellite is limited to a small value andthus, n₁₆ preferably stays not more than 5 times the value, as denotedby the hatching of FIG. 11. The point 104 within the hatching area 103indicates, for example, that the antenna radius r₂ of a two timesgreater value is matched with a 4× value of the transmitting energy.Also, the point 105 represents that the transmission energy should bedoubled when r₂ is about 5× greater. Those values are all within afeasible range.

The value of n₁₆ not greater than 5× value is expressed using A₁ and A₂as:n ₁₆=((A ₁ +A ₂)/A ₁)²≦5Hence, A₂≦1.23A₁.

If the distance between any two signal point group segments shown inFIG. 10 is 2A(4) and the maximum amplitude is 2A(16), A(4) andA(16)-A(4) are proportional to A₁ and A₂ respectively. Hence,(A(16))²≦5(A(14))² is established.

The action of a modified 64 ASPK transmission will be described as thethird receiver 43 can perform 64-state QAM demodulation.

FIG. 12 is a vector diagram in which each signal point group segmentcontains 16 signal points as compared with 4 signal points of FIG. 10.The first signal point group segment 91 in FIG. 12 has a 4×4 matrix of16 signal points allocated at equal intervals including the point 170.For providing compatibility with 4 PSK, A₁≧A_(T0) has to be satisfied.If the radius of the antenna 42 of the third receiver 43 is r₃ and thetransmitting energy is n₆₄, the equation is expressed as:r ₃ ²={6²/(n−1)}r ₁ ²

This relation between r₃ and n of a 64 QAM signal is also shown in thegraphic representation of FIG. 13.

It is under stood that the signal point assignment shown in FIG. 12allows the second receiver 33 to demodulate only two-bit patterns of 4PSK data. Hence, it is desired for having compatibility between thefirst, second, and third receivers that the second receiver 33 isarranged capable of demodulating a modified 16 QAM form from the 64 QAMmodulated signal.

The compatibility between the three discrete receivers can beimplemented by three-level grouping of signal points, as illustrated inFIG. 14. The description will be made referring to the first quadrant inwhich the first signal point group segment 91 represents the two-bitpattern 11 of the first data stream.

In particular, a first sub segment 181 in the first signal point groupsegment 91 is assigned the two-bit pattern 11 of the second data stream.Equally, a second 182, a third 183, and a fourth sub segment 184 areassigned 01, 00, and 10 of the same respectively. This assignment isidentical to that shown in FIG. 7.

The signal point allocation of the third data stream will now beexplained referring to the vector diagram of FIG. 15 which shows thefirst quadrant. As shown, the four signal points 201, 205, 209, 213represent the two-bit pattern of 11, the signal points 202, 206, 210,214 represent 01, the signal points 203, 207, 211, 215 represent 00, andsignal points 204, 208, 212, 216 represent 10. Accordingly, the two-bitpatterns of the third data stream can be transmitted separately of thefirst and second data streams. In other words, two-bit data of the threedifferent signal levels can be transmitted respectively.

As understood, the present invention permits not only transmission ofsix-bit data but also interception of three, two-bit, four-bit, andsix-bit, different bit length data with their respective receivers whilethe signal compatibility remains between three levels.

The signal point allocation for providing compatibility between thethree levels will be described.

As shown in FIG. 15, A₁≧A_(T0) is essential for allowing the firstreceiver 23 to receive the first data stream.

It is needed to space any two signal points from each other by such adistance that the sub segment signal points, e.g. 182, 183, 184, of thesecond data stream shown in FIG. 15 can be distinguished from the signalpoint 91 shown in FIG. 10.

FIG. 15 shows that they are spaced by ⅔A₂. In this case, the distancebetween the two signal points 201 and 202 in the first sub segment 181is A₂/6. The transmitting energy needed for signal interception with thethird receiver 43 is now calculated. If the radius of the antenna 32 isr₃ and the needed transmitting energy is n₆₄ times the 4 PSKtransmitting energy, the equation is expressed as:r ₃ ²=(12r ₁)²/(n−1)This relation is also denoted by the curve 211 in FIG. 16. For example,if the transmitting energy is 6 or 9 times greater than that for 4 PSKtransmission at the point 223 or 222, the antenna 32 having a radius of8× or 6× value respectively can intercept the first, second, and thirddata streams for demodulation. As the signal point distance of thesecond data stream is close to ⅔A₂, the relation between r₁ and r₂ isexpressed by:r ₂ ²=(3r ₁)²/(n−1)Therefore, the antenna 32 of the second receiver 33 has to be a littlebit increased in radius as denoted by the curve 223.

As understood, while the first and second data streams are transmittedtrough a traditional satellite which provides a small signaltransmitting energy, the third data stream can also be transmittedthrough a future satellite which provides a greater signal transmittingenergy without interrupting the action of the first and second receivers23, 33 or with no need of modification of the same and thus, both thecompatibility and the advancement will highly be ensured.

The signal receiving action of the second receiver 33 will first bedescribed. As compared with the first receiver 23 arranged forinterception with a small radius r₁ antenna and demodulation of the 4PSK modulated signal of the digital transmitter 51 or the first datastream of the signal of the transmitter 1, the second receiver 33 isadopted for perfectly demodulating the 16 signal state two-bit data,shown in FIG. 10, or second data stream of the 16 QAM signal from thetransmitter 1. In total, four-bit data including also the first datastream can be demodulated. The ratio between A₁ and A₂ is howeverdifferent in the two transmitters. The two different data are loaded toa demodulation controller 231 of the second receiver 33, shown in FIG.21, which in turn supplies their respective threshold values to thedemodulating circuit for AM demodulation.

The block diagram of the second receiver 33 in FIG. 21 is similar inbasic construction to that of the first receiver 23 shown in FIG. 19.The difference is that the radius r₂ of the antenna 32 is greater thanr₁ of the antenna 22. This allows the second receiver 33 to identify asignal component involving a smaller signal point distance. Thedemodulator 35 of the second receiver 33 also contains a first 232 and asecond data stream reproducing unit 233 in addition to the demodulationcontroller 231. There is provided a first discrimination/reproductioncircuit 136 for AM demodulation of modified 16 QAM signals. Asunderstood, each carrier is a four-bit signal having two, positive andnegative, threshold values about the zero level. As apparent from thevector diagram, of FIG. 22, the threshold values are varied depending onthe transmitting energy of a transmitter since the transmitting signalof the embodiment is a modified 16 QAM signal. When the referencethreshold is TH₁₆, it is determined by, as shown in FIG. 22:TH ₁₆=(A ₁ +A ₂/2)/(A ₁ +A ₂)

The various data for demodulation including A₁ and A₂ or TH₁₆, and thevalue m for multiple-bit modulation are also transmitted from thetransmitter 1 as carried in the first data stream. The demodulationcontroller 231 may be arranged for recovering such demodulation datathrough statistic process of the received signal.

A way of determining the shift factor A₁/A₂ will be described withreference to FIG. 26. A change of the shift factor A₁/A₂ causes a changeof the threshold value. Increase of a difference of a value of A₁/A₂ setat the receiver side from a value of A₁/A₂ set at the transmitter sidewill increase the error rate. Referring to FIG. 26, the demodulatedsignal from the second data stream reproducing unit 233 may be fed backto the demodulation controller 231 to change the shift factor A₁/A₂ in adirection to increase the error rate. By this arrangement, the thirdreceiver 43 may not demodulate the shift factor A₁/A₂, so that thecircuit construction can be simplified. Further, the transmitter may nottransmit the shift factor A₁/A₂, so that the transmission capacity canbe increased. This technique can be applied also to the second receiver33.

FIGS. 25(a) and 25(b) are views showing signal point allocations for theC-CDM signal points, wherein signal points are added by shifting in thepolar coordinate direction (r, θ). The previously described C-CDM ischaracterized in that the signal points are shifted in the rectangularcoordinate direction, i.e. XY direction; therefore it is referred to asrectangular coordinate system C-CDM. Meanwhile, this C-CDM characterizedby the shifting of signal points in the polar coordinate direction, i.e.r, θ direction, is referred to as polar coordinate system C-CDM.

FIG. 25(a) shows the signal allocation of 8PS-APSK signals, wherein foursignal points are added by shifting each of 4 QPSK signals in the radiusr direction of the polar coordinate system. In this manner, the APSK ofpolar coordinate system C-CDM having 8 signal points is obtained fromthe QPSK as shown in FIG. 25(a). As the pole is shifted in the polarcoordinate system to add signal points in this APSK, it is referred toas shifted pole-APSK, i.e SP-APSK in the abbreviated form. In this case,coordinate value of the newly added four QPSK signals 85 are specifiedby using a shift factor S₁ as shown in FIG. 139. Namely, 8PS-APSK signalpoints includes an ordinary QPSK signal points 83 (r₀, θ₀) and a signalpoint ((S₁+1)r₀, θ₀) obtained by shifting the signal point 83 in theradius r direction by an amount of S₁r₀. Thus, a 1-bit subchannel 2 isobtained in addition to a 2-bit subchannel 1 identical with the QPSK.

Furthermore, as shown in the constellation diagram of FIG. 140, neweight signal points, represented by coordinates (r₀+S₂r₀, θ₀) and(r₀+S₁r₀+S₂r₀, θ₀), can be added by shifting the eight signal points(r₀, θ₀) and (r₀+S₁r₀, θ₀) in the radius r direction. As this allows twokinds of allocations, a 1-bit subchannel is obtained and is referred toas 16PS-APSK which provides the 2-bit subchannel 1, 1-bit subchannel 2,and 1-bit subchannel 3. As the 16-PS-APSK disposes the signal points onthe lines of θ=¼·(2n+1)π, it allows the ordinary QPSK receiver explainedwith reference to FIG. 19 to reproduce the carrier wave to demodulatethe first subchannel of 2-bit although the second subchannel cannot bedemodulated. As described above, the C-CDM method of shifting the signalpoints in the polar coordinate direction is useful in expanding thecapacity of information data transmission while assuring compatibilityto the PSK, especially to the QPSK receiver, a main receiver for thepresent satellite broadcast service. Therefore, without losing the firstgeneration viewers of the satellite broadcast service based on the PSK,the broadcast service will advance to a second generation stage whereinthe APSK will be used to increase transmittable information amount byuse of the multi-level modulation while maintaining compatibility.

In FIG. 25(b), the signal points are allocated on the lines of θ=π/8.With this arrangement, the 16 PSK signal points are reduced or limitedto 12 signal points, i.e. 3 signal points in each quadrant. With thislimitation, these three signal points in each quadrant are roughlyregarded as one signal point for 4 QPSK signals. Therefore, this enablesthe QPSK receiver to reproduce the first subchannel in the same manneras in the previous embodiment.

More specifically, the signal points are disposed on the lines of θ=π/4,θ=π/4+π/8, and θ=π/4−π/8. In other words, the added signals are offsetby an amount ±θ in the angular direction of the polar coordinate systemfrom the QPSK signals disposed on the lines of θ=π/4. Since all thesignals are in the range of θ=π/4±π/8, they can be regarded as one ofQPSK signal points on the line of θ=π/4. Although the error rate islowered a little bit in this case, the QPSK receiver 23 shown in FIG. 19can discriminate these points as four signal points angularly allocated.Thus, 2-bit data can be reproduced.

In case of the angular shift C-CDM, if signal points are disposed on thelines of π/n, the carrier wave reproduction circuit can reproduce thecarrier wave by the use of an n-multiplier circuit in the same manner asin other embodiments. If the signal points are not disposed on the linesof π/n, the carrier wave can be reproduced by transmitting severalcarrier information within a predetermined period in the same manner asin other embodiment.

Assuming that an angle between two signal points of the QPSK or8-SP-APSK is 2θ₀ in the polar coordinate system and a first angularshift factor is P1, two signal points (r₀, θ₀+P₁θ₀) and (r₀, θ₀−P₁θ₀)are obtained by shifting the QPSK signal point in the angular θdirection by an amount ±P₁θ₀. Thus, the number of signal points aredoubled. Thus, the 1-bit subchannel 3 can be added and is referred to as8-SP-PSK of P=P1. If eight signal points are further added by shiftingthe 8-SP-PSK signals in the radius r direction by an amount S₁r₀, itwill become possible to obtain 16-SP-APSK (P, S₁ type) as shown in FIG.142. The subchannels 1 and 2 can be reproduced by two 8PS-PSKs havingthe same phase with each other. Returning to FIG. 25(b), as the C-CDMbased on the angular shift in the polar coordinate system can be appliedto the PSK as shown in FIG. 141, this will be adopted to the firstgeneration satellite broadcast service. However, if adopted to thesecond generation satellite broadcasting based on the APSK, this polarcoordinate system C-CDM is inferior in that signal points in the samegroup cannot be uniformly spaced as shown in FIG. 142. Accordingly,utilization efficiency of electric power is worsened. On the other hand,the rectangular coordinate system C-CDM has good compatibility to thePSK.

The system shown in FIG. 25(b) is compatible with both the rectangularand polar coordinate systems. As the signal points are disposed on theangular lines of the 16 PSK, they can be demodulated by the 16 PSK.Furthermore, as the signal points are divided into groups, the QPSKreceiver can be used for demodulation. Still further, as the signalpoints are also allocated to suit for the rectangular coordinate system,the demodulation will be performed by the 16-SRQAM. Consequently, thecompatibility between the rectangular coordinate system C-CDM and thepolar coordinate system C-CDM can be assured in any of the QPSK, 16PSK,and 16-SRQAM.

The demodulation controller 231 has a memory 231 a for storing thereindifferent threshold values (i.e., the shift factors, the number ofsignal points, the synchronization rules, etc.) which correspond todifferent channels of TV broadcast. When receiving one of the channelsagain, the values corresponding to the receiving channel will be readout of the memory to thereby stabilize the reception quickly.

If the demodulation data is lost, the demodulation of the second datastream will hardly be executed. This will be explained referring to aflow chart shown in FIG. 24.

Even if the demodulation data is not available, demodulation of the 4PSK at Step 313 and of the first data stream at Step 301 can beimplemented. At Step 302, the demodulation data retrieved by the firstdata stream reproducing unit 232 is transferred to the demodulationcontroller 231. If m is 4 or 2 at Step 303, the demodulation controller231 triggers demodulation of 4 PSK or 2 PSK at Step 313. If not, theprocedure moves to Step 310. At Step 305, two threshold values TH₈ andTH₁₆ are calculated. The threshold value TH₁₆ for AM demodulation is fedat Step 306 from the demodulation controller 231 to both the first 136and the second discrimination/reproduction circuit 137.

Hence, demodulation of the modified 16 QAM signal and reproduction ofthe second data stream can be carried out at Steps 307 and 315respectively. At Step 308, the error rate is examined and if high, theprocedure returns to Step 313 for repeating the 4 PSK demodulation.

As shown in FIG. 22, the signal points 85, 83, are aligned on a line atan angle of cos(ωt+nπ/2) while 84 and 86 are off the line. Hence, thefeedback of a second data stream transmitting carrier wave data from thesecond data stream reproducing unit 233 to a carrier reproducing circuit131 is carried out so that no carrier needs to be extracted at thetiming of the signal points 84 and 86.

The transmitter 1 is arranged to transmit carrier timing signals atintervals of a given time with the first data stream for the purpose ofcompensation for no demodulation of the second data stream. The carriertiming signal enables to identify the signal points 83 and 85 of thefirst data stream regardless of demodulation of the second data stream.Hence, the reproduction of carrier wave can be triggered by thetransmitting carrier data to the carrier reproducing circuit 131.

It is then examined at Step 304 of the flow chart of FIG. 24 whether mis 16 or not upon receipt of such a modified 64 QAM signal as shown inFIG. 23. At Step 310, it is also examined whether m is more than 64 ornot. If it is determined at Step 311 that the received signal has noequal distance signal point constellation, the procedure goes to Step312. The signal point distance TH₆₄ of the modified 64 QAM signal iscalculated from:TH ₆₄=(A ₁ +A ₂/2)/(A ₁ +A ₂)This calculation is equivalent to that of TH₁₆ but its resultantdistance between signal points is smaller.

If the signal point distance in the first sub segment 181 is A₃, thedistance between the first 181 and the second sub segment 182 isexpressed by (A₂−2A₃). Then, the average distance is (A₂−2A₃)/(A₁+A₂)which is designated as d₆₄. When d₆₄ is smaller than T₂ which representsthe signal point discrimination capability of the second receiver 33,any two signal points in the segment will hardly be distinguished fromeach other. This judgement is executed at Step 313. If d₆₄ is out of apermissive range, the procedure moves back to Step 313 for 4 PSK modedemodulation. If d₆₄ is within the range, the procedure advances to Step305 for allowing the demodulation of 16 QAM at Step 307. If it isdetermined at Step 308 that the error rate is too high, the proceduregoes back to Step 313 for 4 PSK mode demodulation.

When the transmitter 1 supplied a modified 8 QAM signal such as shown inFIG. 25(a) in which all the signal points are at angles ofcos(2πf+n·π/4), the carrier waves of the signal are lengthened to thesame phase and will thus be reproduced with much ease. At the time,two-bit data of the first data stream are demodulated with the 4-PSKreceiver while one-bit data of the second data stream is demodulatedwith the second receiver 33 and the total of three-bit data can bereproduced.

The third receiver 43 will be described in more detail. FIG. 26 shows ablock diagram of the third receiver 43 similar to that of the secondreceiver 33 in FIG. 21. The difference is that a third data streamreproducing unit 234 is added and also, the discrimination/reproductioncircuit has a capability of identifying eight-bit data. The antenna 42of the third receiver 43 has a radius r₃ greater than r₂ thus allowingsmaller distance state signals, e.g. 32- or 64-state QAM signals, to bedemodulated. For demodulation of the 64 QAM signal, the firstdiscrimination/reproduction circuit 136 has to identify 8 digital levelsof the detected signal in which seven different threshold levels areinvolved. As one of the threshold values is zero, three are contained inthe first quadrant.

FIG. 27 shows a space diagram of the signal in which the first quadrantcontains three different threshold values.

As shown in FIG. 27, when the three normalized threshold values are TH1₆₄, TH2 ₆₄, and TH3 ₆₄, they are expressed by:TH1₆₄=(A ₁ +A ₃/2)/(A ₁ +A ₂)TH2₆₄=(A ₁ +A ₂/2)/(A ₁ +A ₂) andTH3₆₄=(A ₁ +A ₂ −A ₃/2)/(A ₁ +A ₂).

Through AM demodulation of a phase detected signal using the threethreshold values, the third data stream can be reproduced like the firstand second data stream explained with FIG. 21. The third data streamcontains e.g. four signal points 201, 202, 203, 204 at the first subsegment 181 shown in FIG. 23 which represent 4 values of two-bitpattern. Hence, six digits or modified 64 QAM signals can bedemodulated.

The demodulation controller 231 detects the value m, A₁, A₂, and A₃ fromthe demodulation data contained in the first data stream demodulated atthe first data stream reproducing, unit 232 and calculates the threethreshold values TH1 ₆₄, TH2 ₆₄, and TH3 ₆₄ which are then fed to thefirst 136 and the second discrimination/reproduction circuit 137 so thatthe modified 64 QAM signal is demodulated with certainty. Also, if thedemodulation data have been scrambled, the modified 64 QAM signal can bedemodulated only with a specific or subscriber receiver. FIG. 28 is aflow chart showing the action of the demodulation controller 231 formodified 64 QAM signals. The difference from the flow chart fordemodulation of 16 QAM shown in FIG. 24 will be explained. The proceduremoves from Step 304 to Step 320 where it is examined whether m=32 ornot. If m=32, demodulation of 32 QAM signals is executed at Step 322. Ifnot, the procedure moves to Step 321 where it is examined whether m=64or not. If yes, A₃ is examined at Step 323. If A₃ is smaller than apredetermined value, the procedure moves to Step 305 and the samesequence as of FIG. 24 is implemented. If it is judged at Step 323 thatA₃ is not smaller than the predetermined value, the procedure goes toStep 324 where the threshold values are calculated. At Step 325, thecalculated threshold values are fed to the first and seconddiscrimination/reproduction circuits and at Step 326, the demodulationof the modified 64 QAM signal is carried out. Then, the first, second,and third data streams are reproduced at Step 327. At Step 328, theerror rate is examined. If the error rate is high, the procedure movesto Step 305 where the 16 QAM demodulation is repeated and if low, thedemodulation of the 64 QAM is continued.

The action of carrier wave reproduction needed for execution of asatisfactory demodulating procedure will now be described. The scope ofthe present invention includes reproduction of the first data stream ofa modified 16 or 64 QAM signal with the use of a 4 PSK receiver.However, a common 4 PSK receiver rarely reconstructs carrier waves, thusfailing to perform a correct demodulation. For compensation, somearrangements are necessary at both the transmitter and receiver sides.

Two techniques for the compensation are provided according to thepresent invention. A first technique relates to transmission of signalpoints aligned at angles of (2n−1)π/4 at intervals of a given time. Asecond technique offers transmission of signal points arranged atintervals of an angle of n π/8.

According to the first technique, the eight signal points including 83and 85 are aligned at angles of π/4, 3π/4, 5π/4, and 7π/4, as shown inFIG. 38. In action, at least one of the eight signal points istransmitted during sync time slot periods 452, 453, 454, 455 arranged atequal intervals of a time in a time slot gap 451 shown in the time chartof FIG. 38. Any desired signal points are transmitted during the othertime slots. The transmitter 1 is also arranged to assign a data for thetime slot interval to the sync timing data region 499 of a sync datablock, as shown in FIG. 41.

The content of a transmitting signal will be explained in more detailreferring to FIG. 41. The time slot group 451 containing the sync timeslots 452, 453, 454, 455 represents a unit data stream or block 491carrying a data of Dn.

The sync time slots in the signal are arranged at equal intervals of agiven time determined by the time slot interval or sync timing data.Hence, when the arrangement of the sync time slots is detected,reproduction of carrier waves will be executed slot by slot throughextracting the sync timing data from their respective time slots. Such async timing data S is contained in a sync block 493 accompanied at thefront end of a data frame 492, which is consisted of a number of thesync time slots denoted by the hatching in FIG. 41. Accordingly, thedata to be extracted for carrier wave reproduction are increased, thusallowing the 4 PSK receiver to reproduce desired carrier waves at higheraccuracy and efficiency.

The sync block 493 comprises sync data regions 496, 497, 498, - - -containing sync data S1, S2, S3, - - - respectively which include uniquewords and demodulation data. The phase sync signal assignment region 499is accompanied at the end of the sync block 493, which holds a data ofI_(T) including information about interval arrangement and assignment ofthe sync time slots.

The signal point data in the phase sync time slot has a particular phaseand can thus be reproduced by the 4 PSK receiver. Accordingly, I_(T) inthe phase sync signal assignment region 499 can be retrieved withouterror thus ensuring the reproduction of carrier waves at accuracy.

As shown in FIG. 41, the sync block 493 is followed by a demodulationdata block 501 which contains demodulation data about threshold voltagesneeded for demodulation of the modified multiple-bit QAM signal. Thisdata is essential for demodulation of the multiple-bit QAM signal andmay preferably be contained in a region 502 which is a part of the syncblock 493 for ease of retrieval.

FIG. 42 shows the assignment of signal data for transmission of burstform signals through a TDMA method.

The assignment is distinguished from that of FIG. 41 by the fact that aguard period 521 is inserted between any two adjacent Dn data blocks491, 491 for interruption of the signal transmission. Also, each datablock 491 is accompanied at front end a sync region 522 thus forming adata block 492. During the sync region 522, the signal points at a phaseof (2n−1)π/4 are only transmitted. Accordingly, the carrier wavereproduction will be feasible with the 4 PSK receiver. Morespecifically, the sync signal and carrier waves can be reproducedthrough the TDMA method.

The carrier wave reproduction of the first receiver 23 shown in FIG. 19will be explained in more detail referring to FIGS. 43 and 44. As shownin FIG. 43, an input signal is fed through the input unit 24 to a syncdetector circuit 541 where it is sync detected. A demodulated signalfrom the sync detector 541 is transferred to an output circuit 542 forreproduction of the first data stream. A data of the phase sync signalassignment data region 499 (shown in FIG. 41) is retrieved with anextracting timing controller circuit 543 so that the timing of syncsignals of (2n−1)π/4 data can be acknowledged and transferred as a phasesync control pulse 561 shown in FIG. 44 to a carrier reproductioncontrolling circuit 544. Also, the demodulated signal of the syncdetector circuit 541 is fed to a frequency multiplier circuit 545 whereit is 4× multiplied prior to transmitted to the carrier reproductioncontrolling circuit 544. The resultant signal denoted by 562 in FIG. 44contains a true phase data 563 and other data. As illustrated in a timechart 564 of FIG. 44, the phase sync time slots 452 carrying the(2n−1)π/4 data are also contained at equal intervals. At the carrierreproducing controlling circuit 544, the signal 562 is sampled by thephase sync control pulse 561 to produce a phase sample signal 565 whichis then converted through sample-hold action to a phase signal 566. Thephase signal 566 of the carrier reproduction controlling circuit 544 isfed across a loop filter 546 to a VCO 547 where its relevant carrierwave is reproduced. The reproduced carrier is then sent to the syncdetector circuit 541.

In this manner, the signal point data of the (2n−1)π/4 phase denoted bythe shaded areas in FIG. 39 is recovered and utilized so that a correctcarrier wave can be reproduced by 4× or 16× frequency multiplication.Although a plurality of phases are reproduced at the time, the absolutephases of the carrier can be successfully be identified with the used ofa unique word assigned to the sync region 496 shown in FIG. 41.

For transmission of a modified 64 QAM signal such as shown in FIG. 40,signal points in the phase sync areas 471 at the (2n−1)π/4 phase denotedby the hatching are assigned to the sync time slots 452, 452 b, etc. Itscarrier can be reproduced hardly with a common 4 PSK receiver butsuccessfully with the first receiver 23 of 4 PSK mode provided with thecarrier reproducing circuit of the embodiment.

The foregoing carrier reproducing circuit is of COSTAS type. A carrierreproducing circuit of reverse modulation type will now be explainedaccording to the embodiment.

FIG. 45 shows a reverse modulation type carrier reproducing circuitaccording to the present invention, in which a received signal is fedfrom the input unit 24 to a sync detector circuit 541 for producing ademodulated signal. Also, the input signal is delayed by a first delaycircuit 591 to a delay signal. The delay signal is then transferred to aquadrature phase modulator circuit 592 where it is reverse demodulatedby the demodulated signal from the sync detector circuit 541 to acarrier signal. The carrier signal is fed through a carrier reproductioncontroller circuit 544 to a phase comparator 593. A carrier waveproduced by a VCO 547 is delayed by a second delay circuit 594 to adelay signal which is also fed to the phase comparator 593. At the phasecomparator 594, the reverse demodulated carrier signal is compared inphase with the delay signal thus producing a phase difference signal.The phase difference signal sent through a loop filter 546 to the VCO547 which in turn produces a carrier wave arranged in phase with thereceived carrier wave. In the same manner as of the COSTAS carrierreproducing circuit shown in FIG. 43, an extracting timing controllercircuit 543 performs sampling of signal points contained in the hatchingareas of FIG. 39. Accordingly, the carrier wave of a 16 or 64 QAM signalcan be reproduced with the 4 PSK demodulator of the first receiver 23.

The reproduction of a carrier wave by 16× frequency multiplication willbe explained. The transmitter 1 shown in FIG. 1 is arranged to modulateand transmit a modified 16 QAM signal with assignment of its signalpoints at nπ/8 phase as shown in FIG. 46. At the first receiver 23 shownin FIG. 19, the carrier wave can be reproduced with its COSTAS carrierreproduction controller circuit containing a 16× multiplier circuit 661shown in FIG. 48. The signal points at each nπ/8 phase shown in FIG. 46are processed at the first quadrant b the action of the 16× multipliercircuit 661, whereby the carrier will be reproduced by the combinationof a loop filter 546 and a VCO 541. Also, the absolute phase may bedetermined from 16 different phases by assigning a unique word to thesync region.

The arrangement of the 16× multiplier circuit will be explainedreferring to FIG. 48. A sum signal and a difference signal are producedfrom the demodulated signal by an adder circuit 662 and a subtractorcircuit 663 respectively and then, multiplied each other by a multiplier664 to a cos 2θ signal. Also, a multiplier 665 produces a sin 2θ signal.The two signals are then multiplied by a multiplier 666 to a sin 4θsignal.

Similarly, a sin 8θ signal is produced from the two, sin 2θ and cos 2θ,signals by the combination of an adder circuit 667, a subtracter circuit668, and a multiplier 670. Furthermore, a sin 16θ signal is produced bythe combination of an adder circuit 671, a subtractor circuit 672, and amultiplier 673. Then, the 16× multiplication is completed.

Through the foregoing 16× multiplication, the carrier wave of all thesignal points of the modified 16 QAM signal shown in FIG. 46 willsuccessfully be reproduced without extracting particular signal points.

However, reproduction of the carrier wave of the modified 64 QAM signalshown in FIG. 47 can involve an increase in the error rate due todislocation of some signal points from the sync areas 471.

Two techniques are known for compensation for the consequences. One isinhibiting transmission of the signal points dislocated from the syncareas. This causes the total amount of transmitted data to be reducedbut allows the arrangement to be facilitated. The other is providing thesync time slots as described in FIG. 38. In more particular, the signalpoints in the nπ/8 sync phase areas, e.g. 471 and 471 a, are transmittedduring the period of the corresponding sync time slots in the time slotgroup 451. This triggers an accurate synchronizing action during theperiod thus minimizing phase error.

As now understood, the 16× multiplication allows the simple 4 PSKreceiver to reproduce the carrier wave of a modified 16 or 64 QAMsignal. Also, the insertion of the sync time slots causes the phasicaccuracy to be increased during the reproduction of carrier waves from amodified 64 QAM signal.

As set forth above, the signal transmission system of the presentinvention is capable of transmitting a plurality of data on a singlecarrier wave simultaneously in the multiple signal level arrangement.

More specifically, three different level receivers which have discretecharacteristics of signal intercepting sensitivity and demodulatingcapability are provided in relation to one single transmitter so thatany one of them can be selected depending on a wanted data size to bedemodulated which is proportional to the price. When the first receiverof low resolution quality and low price is acquired together with asmall antenna, its owner can intercept and reproduce the first datastream of a transmission signal. When the second receiver of mediumresolution quality and medium price is acquired together with a mediumantenna, its owner can intercept and reproduce both the first and seconddata streams of the signal. When the third receiver of high resolutionquality and high price is acquired with a large antenna, its owner canintercept and reproduce all the first, second, and third data streams ofthe signal.

If the first receiver is a home-use digital satellite broadcast receiverof low price, it will overwhelmingly be welcome by a majority ofviewers. The second receiver accompanied with the medium antenna costsmore and will be accepted by not common viewers but particular peoplewho wants to enjoy HDTV services. The third receiver accompanied withthe large antenna at least before the satellite output is increased, isnot appropriated for home use and will possibly be used in relevantindustries. For example, the third data stream carrying super HDTVsignals is transmitted via a satellite to subscriber cinemas which canthus play video tapes rather than traditional movie films and run moviesbusiness at low cost.

When the present invention is applied to a TV signal transmissionservice, three different quality pictures are carried on one signalchannel wave and will offer compatibility with each other. Although thefirst embodiment refers to a 4 PSK, a modified 8 QAM, a modified 16 QAM,and a modified 64 QAM signal, other signals will also be employed withequal success including a 32 QAM, a 256 QAM, an 8 PSK, a 16 PSK, a 32PSK signal. It would be understood that the present invention is notlimited to a satellite transmission system and will be applied to aterrestrial communications system or a cable transmission system.

Embodiment 2

A second embodiment of the present invention is featured in which thephysical multi-level arrangement of the first embodiment is divided intosmall levels through e.g. discrimination in error correction capability,thus forming a logic multi-level construction. In the first embodiment,each multi-level channel has different levels in the electric signalamplitude or physical demodulating capability. The second embodimentoffers different levels in the logic reproduction capability such aserror correction. For example, the data D₁ in a multi-level channel isdivided into two, D₁₋₁ and D₁₋₂, components and D₁₋₁ is more increasedin the error correction capability than D₁₋₂ for discrimination.Accordingly, as the error detection and correction capability isdifferent between D₁₋₁ and D₁₋₂ at demodulation, D₁₋₁ can successfullybe reproduced within a given error rate when the C/N level of anoriginal transmitting signal is as low as disenabling the reproductionof D₁₋₂. This will be implemented using the logic multi-levelarrangement.

More specifically, the logic multi-level arrangement is consisted ofdividing data of a modulated multi-level channel and discriminatingdistances between error correction codes by mixing error correctioncodes with product codes for varying error correction capability. Hence,a more multi-level signal can be transmitted.

In fact, a D₁ channel is divided into two sub channels D₁₋₁ and D₁₋₂ anda D₂ channel is divided into two sub channels D₂₋₁ and D₂₋₂.

This will be explained in more detail referring to FIG. 87 in which D₁₋₁is reproduced from a lowest C/N signal. If the C/N rate is d at minimum,three components D₁₋₂, D₂₋₁ and D₂₋₂ cannot be reproduced while D₁₋₁ isreproduced. If C/N is not less than c, D₁₋₂ can also be reproduced.Equally, when C/N is b, D₂₋₁ is reproduced and when C/N is a, D₂₋₂ isreproduced. As the C/N rate increases, the reproducible signal levelsare increased in number. The lower the C/N, the fewer the reproduciblesignal levels. This will be explained in the form of relation betweentransmitting distance and reproducible C/N value referring to FIG. 86.In common, the C/N value of a received signal is decreased in proportionto the distance of transmission as expressed by the real line 861 inFIG. 86. It is now assumed that the distance from a transmitter antennato a receiver antenna is La when C/N=a, Lb when C/N=b, Lc when C/N=c, Ldwhen C/N=d, and Le when C/N=e. If the distance from the transmitterantenna is greater than Ld, D₁₋₁ can be reproduced as shown in FIG. 85where the receivable area 862 is denoted by the hatching. In otherwords, D₁₋₁ can be reproduced within a most extended area. Similarly,D₁₋₂ can be reproduced in an area 863 when the distance is not more thanLc. In this area 863 containing the area 862, D₁₋₁ can with no doubt bereproduced. In a small area 854, D₂₋₁ can be reproduced and in asmallest area 865, D₂₋₂ can be reproduced. As understood, the differentdata levels of a channel can be reproduced corresponding to degrees ofdeclination in the C/N rate. The logic multi-level arrangement of thesignal transmission system of the present invention can provide the sameeffect as of a traditional analogue transmission system in which theamount of receivable data is gradually lowered as the C/N ratedecreases.

The construction of the logic multi-level arrangement will be describedin which there are provided two physical levels and two logic levels.FIG. 87 is a block diagram of a transmitter 1 which is substantiallyidentical in construction to that shown in FIG. 2 and describedpreviously in the first embodiment and will no further be explained indetail. The only difference is that error correction code encoders areadded as abbreviated to ECC encoders. The divider circuit 3 has fouroutputs 1-1, 1-2, 2-1, and 2-2 through which four signals D₁₋₁, D₁₋₂,D₂₋₁, and D₂₋₂ divided from an input signal are delivered. The twosignals D₁₋₁ and D₁₋₂ are fed to two, main and sub, ECC encoders 872 a,873 a of a first ECC encoder 871 a respectively for converting to errorcorrection code forms.

The main ECC encoder 872 a has a higher error correction capability thanthat of the sub ECC encoder 873 a. Hence, D₁₋₁ can be reproduced at alower rate of C/N than D₁₋₂ as apparent from the CN-level diagram ofFIG. 85. More particularly, the logic level of D₁₋₁ is less affected bydeclination of the C/N than that of D₁₋₂. After error correction codeencoding, D₁₋₁ and D₁₋₂ are summed by a summer 874 a to a D₁ signalwhich is then transferred to the modulator 4. The other two signals D₂₋₁and D₂₋₂ of the divider circuit 3 are error correction encoded by two,main and sub, ECC encoders 872 b, 873 b of a second ECC encoder 871 brespectively and then, summed by a summer 874 b to a D₂ signal which istransmitted to the modulator 4. The main ECC encoder 872 b is higher inthe error correction capability than the sub ECC encoder 873 b. Themodulator 4 in turn produces from the two, D₁ and D₂, input signals amulti-level modulated signal which is further transmitted from thetransmitter unit 5. As understood, the output signal from thetransmitter 1 has two physical levels D₁ and D₂ and also, four logiclevels D₁₋₁, D₁₋₂, D₂₋₁, and D₂₋₂ based on the two physical levels forproviding different error correction capabilities.

The reception of such a multi-level signal will be explained. FIG. 88 isa block diagram of a second receiver 33 which is almost identical inconstruction to that shown in FIG. 21 and described in the firstembodiment. The second receiver 33 arranged for intercepting multi-levelsignals from the transmitter 1 shown in FIG. 87 further comprises afirst 876 a and a second ECC decoder 876 b, in which the demodulation ofQAM, or any of ASK, PSK, and FSK if desired, is executed.

As shown in FIG. 88, a receiver signal is demodulated by the demodulator35 to the two, D₁ and D₂, signals which are then fed to two dividers 3 aand 3 b respectively where they are divided into four logic levels D₁₋₁,D₁₋₂, D₂₋₁, and D₂₋₂. The four signals are transferred to the first 876a and the second ECC decoder 876 b in which D₁₋₁ is error corrected by amain ECC decoder 877 a, D₁₋₂ by a sub ECC decoder 878 a, D₂₋₁ by a mainECC decoder 877 b, D₂₋₂ by a sub ECC decoder 878 b before all sent tothe summer 37. At the summer 37, the four, D₁₋₁, D₁₋₂, D₂₋₁, and D₂₋₂,error corrected signals are summed to a signal which is then deliveredfrom the output unit 36.

Since D₁₋₁ and D₂₋₁ are higher in the error correction capability thanD₁₋₂ and D₂₋₂ respectively, the error rate remains less than a givenvalue although C/N is fairly low as shown in FIG. 85 and thus, anoriginal signal will be reproduced successfully.

The action of discriminating the error correction capability between themain ECC decoders 877 a, 877 b and the sub ECC decoders 878 a, 878 bwill now be described in more detail. It is a good idea for having adifference in the error correction capability to use in the sub ECCdecoder a common coding technique, e.g. Reed-Solomon or BCH method,having a standard code distance and in the main ECC decoder, anotherencoding technique in which the distance between correction codes isincreased using Reed-Solomon codes, their product codes, or otherlong-length codes. A variety of known techniques for increasing theerror correction code distance have been introduced and will no moreexplained. The present invention can be associated with any knowntechnique for having the logic multi-level arrangement.

The logic multi-level arrangement will be explained in conjunction witha diagram of FIG. 89 showing the relation between C/N and error rateafter error correction. As shown, the straight line 881 represents D₁₋₁at the C/N and error rate relation and the line 882 represents D₁₋₂ atsame.

As the C/N rate of an input signal decreases, the error rate increasesafter error correction. If C/N is lower than a given value, the errorrate exceeds a reference value Eth determined by the system designstandards and no original data will normally be reconstructed. When C/Nis lowered to less than e, the D₁ signal fails to be reproduced asexpressed by the line 881 of D₁₋₁ in FIG. 89. When e≦C/N<d, D₁₋₁ of theD₁ signal exhibits a higher error rate than Eth and will not bereproduced.

When C/N is d at the point 885 d, D₁₋₁ having a higher error correctioncapability than D₁₋₂ becomes not higher in the error rate than Eth andcan be reproduced. At the time, the error rate of D₁₋₂, remains higherthan Eth after error correction and will no longer be reproduced.

When C/N is increased up to c at the point 885 c, D₁₋₂ becomes nothigher in the error rate than Eth and can be reproduced. At the time,D₂₋₁ and D₂₋₂ remain in no demodulation state. After the C/N rate isincreased further to b′, the D₂ signal becomes ready to be demodulated.

When C/N is increased to b at the point 885 b, D₂₋₁ of the D₂ signalbecomes not higher in the error rate than Eth and can be reproduced. Atthe time, the error rate of D₂₋₂ remains higher than Eth and will not bereproduced. When C/N is increased up to a at the point 885 a, D₂₋₂becomes not higher than Eth and can be reproduced.

As described above, the four different signal logic levels divided fromtwo, D₁ and D₂, physical levels through discrimination of the errorcorrection capability between the levels, can be transmittedsimultaneously.

Using the logic multi-level arrangement of the present invention inaccompany with a multi-level construction in which at least a part ofthe original signal is reproduced even if data in a higher level islost, digital signal transmission will successfully be executed withoutlosing the advantageous effect of an analogue signal transmission inwhich transmitting data is gradually decreased as the C/N rate becomeslow.

Thanking to up-to-data image data compression techniques, compressedimage data can be transmitted in the logic multi-level arrangement forenabling a receiver station to reproduce a higher quality image thanthat of an analogue system and also, with not sharply but at stepsdeclining the signal level for ensuring signal interception in a widerarea. The present invention can provide an extra effect of themulti-layer arrangement which is hardly implemented by a known digitalsignal transmission system without deteriorating high quality imagedata.

Embodiment 3

A third embodiment of the present invention will be described referringto the relevant drawings.

FIG. 29 is a schematic total view illustrating the third embodiment inthe form of a digital TV broadcasting system. An input video signal 402of super high resolution TV image is fed to an input unit 403 of a firstvideo encoder 401. Then, the signal is divided by a divider circuit 404into three, first, second, and third, data streams which are transmittedto a compressing circuit 405 for data compression before furtherdelivered.

Equally, other three input video signals 406, 407, and 408 are fed to asecond 409, a third 410, and a fourth video encoder 411 respectivelywhich all are arranged identical in construction to the first videoencoder 401 for data compression.

The four first data streams from their respective encoders 401, 409,410, 411 are transferred to a first multiplexer 413 of a multiplexer 412where they are time multiplexed by TDM process to a first data streammultiplex signal which is fed to a transmitter 1.

A part or all of the four second data streams from their respectiveencoders 401, 409, 410, 411 are transferred to a second multiplexer 414of the multiplexer 412 where they are time multiplexed to a second datastream multiplex signal which is then fed to the transmitter 1. Also, apart or all of the four third data streams are transferred to a thirdmultiplexer 415 where they are time multiplexed to a third data streammultiplex signal which is then fed to the transmitter 1.

The transmitter 1 performs modulation of the three data stream signalswith its modulator 4 by the same manner as described in the firstembodiment. The modulated signals are sent from a transmitter unit 5through an antenna 6 and an uplink 7 to a transponder 12 of a satellite10 which in turn transmits it to three different receivers including afirst receiver 23.

The modulated signal transmitted through a downlink 21 is intercepted bya small antenna 22 having a radius r₁ and fed to a first data streamreproducing unit 232 of the first receiver 23 where its first datastream only is demodulated. The demodulated first data stream is thenconverted by a first video decoder 421 to a traditional 425 orwide-picture NTSC or video output signal 426 of low image resolution.

Also, the modulated signal transmitted through a downlink 31 isintercepted by a medium antenna 32 having a radius r₂ and fed to a first232 and a second data stream reproducing unit 233 of a second receiver33 where its first and second data streams are demodulated respectively.The demodulated first and second data streams are then summed andconverted by a second video decoder 422 to an HDTV or video outputsignal 427 of high image resolution and/or to the video output signals425 and 426.

Also, the modulated signal transmitted through a downlink 41 isintercepted by a large antenna 42 having a radius r₃ and fed to a first232, a second 233, and a third data stream reproducing unit 234 of athird receiver 43 where its first, second, and third data streams aredemodulated respectively. The demodulated first, second, and third datastreams are then summed and converted by a third video decoder 423 to asuper HDTV or video output signal 428 of super high image resolution foruse in a video theater or cinema. The video output signals 425, 426, and427 can also be reproduced if desired. A common digital TV signal istransmitted from a conventional digital transmitter 51 and whenintercepted by the first-receiver 23, will be converted to the videooutput signal 426 such as a low resolution NTSC TV signal.

The first video encoder 401 will now be explained in more detailreferring to the block diagram of FIG. 30. An input video signal ofsuper high resolution is fed through the input unit 403 to the dividercircuit 404 where it is divided into four components by sub-band codingprocess. In more particular, the input video signal is separated throughpassing a horizontal lowpass filer 451 and a horizontal highpass filter452 of e.g. QMF mode to two, low and high, horizontal frequencycomponents which are then subsampled to a half of their quantities bytwo subsamplers 453 and 454 respectively. The low horizontal componentis filtered by a vertical lowpass filter 455 and a vertical highpassfilter 456 to a low horizontal low vertical component or H_(L)V_(L)signal and a low horizontal high vertical component or H_(L)V_(H) signalrespectively. The two, H_(L)V_(L) and H_(L)V_(H), signals are thensubsampled to a half by two subsamplers 457 and 458 respectively andtransferred to the compressing circuit 405.

The high horizontal component is filtered by a vertical lowpass filter459 and a vertical highpass filter 460 to a high horizontal low verticalcomponent or H_(H)V_(L) signal and a high horizontal high verticalcomponent or H_(H)V_(H) signal respectively. The two, H_(H)V_(L) andH_(H)V_(H), signals are then subsampled to a half by two subsamplers 461and 462 respectively and transferred to the compressing circuit 405.

H_(L)V_(L) signal is preferably DCT compressed by a first compressor 471of the compressing circuit 405 and fed to a first output 472 as thefirst data stream.

Also, H_(L)V_(H) signal is compressed by a second compressor 473 and fedto a second output 464. H_(H)V_(L) signal is compressed by a thirdcompressor 463 and fed to the second output 464.

H_(H)V_(H) signal is divided by a divider 465 into two, high resolution(H_(H)V_(H)1) and super high resolution (H_(H)V_(H)2), video signalswhich are then transferred to the second output 464 and a third output468 respectively.

The first video decoder 421 will now be explained in more detailreferring to FIG. 31. The first data stream or D₁ signal of the firstreceiver 23 is fed through an input unit 501 to a descrambler 502 of thefirst video decoder 421 where it is descrambled. The descrambled D₁signal is expanded by an expander 503 to H_(L)V_(L) which is then fed toan aspect ratio changing circuit 504. Thus, H_(L)V_(L) signal can bedelivered through an output unit 505 as a standard 500, letterbox format507, wide-screen 508, or sidepanel format NTSC signal 509. The scanningformat may be of non-interlace or interlace type and its NTSC mode linesmay be 525 or doubled to 1050 by double tracing. When the receivedsignal from the digital transmitter 51 is a digital TV signal of 4 PSKmode, it can also be converted by the first receiver 23 and the firstvideo decoder 421 to a TV picture. The second video decoder 422 will beexplained in more detail referring to the block diagram of FIG. 32. TheD₁ signal of the second receiver 33 is fed through a first input 521 toa first expander 522 for data expansion and then, transferred to anoversampler 523 where it is sampled at 2×. The oversampled signal isfiltered by a vertical lowpass filter 524 to H_(L)V_(L). Also, the D₂signal of the second receiver 33 is fed through a second input 530 to adivider 531 where it is divided into three components which are thentransferred to a second 532, a third 533, and a fourth expander 534respectively for data expansion. The three expanded components aresampled at 2× by three oversamplers 535, 536, 537 and filtered by avertical highpass 538, a vertical lowpass 539, and a vertical high-passfilter 540 respectively. Then, H_(L)V_(L) from the vertical lowpassfilter 524 and H_(L)V_(H) from the vertical highpass filter 538 aresummed by an adder 525, sampled by an oversampler 541, and filtered by ahorizontal lowpass filter 542 to a low frequency horizontal videosignal. H_(H)V_(L) from the vertical lowpass filter 539 and H_(H)V_(H)1from the vertical highpass filter 540 are summed by an adder 526,sampled by an oversampler 544, and filtered by a horizontal highpassfilter 545 to a high frequency horizontal video signal. The two, highand low frequency, horizontal video signal are then summed by an adder543 to a high resolution video signal HD which is further transmittedthrough an output unit 546 as a video output 547 of e.g. HDTV format. Ifdesired a traditional NTSC video output can be reconstructed with equalsuccess.

FIG. 33 is a block diagram of the third video decoder 423 in which theD₁ and D₂ signals are fed through a first 521 and a second input 530respectively to a high frequency band video decoder circuit 527 wherethey are converted to an HD signal by the same manner as abovedescribed. The D₃ signal is fed through a third input 551 to a superhigh frequency band video decoder circuit 552 where it is expanded,descrambled, and composed to H_(H)V_(H)2 signal. The HD signal of thehigh frequency band video decoder circuit 527 and the H_(H)V_(H)2 signalof the super high frequency band video decoder circuit 552 are summed bya summer 553 to a super high resolution TV or S-HD signal which is thendelivered through an output unit 554 as a super resolution video output555.

The action of multiplexing in the multiplexer 412 shown in FIG. 29 willbe explained in more detail. FIG. 34 illustrates a data assignment inwhich the three, first, second, and third, data streams D₁, D₂, D₃contain in a period of T six NTSC channel data L1, L2, L3, L4, L5, L6,six HDTV channel data M1, M2, M3, M4, M5, M6 and six S-HDTV channel dataH1, H2, H3, H4, H5, H6 respectively. In action, the NTSC or D₁ signaldata L1 to L6 are time multiplexed by TDM process during the period T.More particularly, H_(L)V_(L) of D₁ is assigned to a domain 601 for thefirst channel. Then, a difference data M1 between HDTV and NTSC or a sumof H_(L)V_(H), H_(H)V_(L), and H_(H)V_(H)1 is assigned to a domain 602for the first channel. Also, a difference data H1 between HDTV and superHDTV or H_(H)V_(H)2 (See FIG. 30) is assigned to a domain 603 for thefirst channel.

The selection of the first channel TV signal will now be described. Whenintercepted by the first receiver 23 with a small antenna coupled to thefirst video decoder 421, the first channel signal is converted to astandard or widescreen NTSC TV signal as shown in FIG. 31. Whenintercepted by the second receiver 33 with a medium antenna coupled tothe second video decoder 422, the signal is converted by summing L1 ofthe first data stream D₁ assigned to the domain 601 and M1 of the seconddata stream D₂ assigned to the domain 602 to an HDTV signal of the firstchannel equivalent in program to the NTSC signal.

When intercepted by the third receiver 43 with a large antenna coupledto the third video decoder 423, the signal is converted by summing L1 ofD₁ assigned to the domain 601, M1 of D₂ assigned to the domain 602, andH₁ of D₃ assigned to the domain 603 to a super HDTV signal of the firstchannel equivalent in program to the NTSC signal. The other channelsignals can be reproduced in an equal manner.

FIG. 35 shows another data assignment L1 of a first channel NTSC signalis assigned to a first domain 601. The domain 601 which is allocated atthe front end of the first data stream D₁, also contains at front a dataS11 including a descrambling data and the demodulation data described inthe first embodiment. A first channel HDTV signal is transmitted as L1and M1. M1 which is thus a difference data between NTSC and HDTV isassigned to two domains 602 and 611 of D₂. If L₁ is a compressed NTSCcomponent of 6 Mbps, M1 is as two times higher as 12 Mbps. Hence, thetotal of L1 and M1 can be demodulated at 18 Mbps with the secondreceiver 33 and the second video decoder 423. According to current datacompression techniques, HDTV compressed signals can be reproduced atabout 15 Mbps. This allows the data assignment shown in FIG. 35 toenable simultaneous reproduction of an NTSC and HDTV first channelsignal. However, this assignment allows no second channel HDTV signal tobe carried. S21 is a descrambling data in the HDTV signal. A firstchannel super HDTV signal component comprises L1, M1, and H1. Thedifference data H1 is assigned to three domains 603, 612, and 613 of D₃.If the NTSC signal is 6 Mbps, the super HDTV is carried at as high as 36Mbps. When a compressed rate is increased, super HDTV video data ofabout 2000 scanning line for reproduction of a cinema size picture forcommercial use can be transmitted with an equal manner.

FIG. 36 shows a further data assignment in which H1 of a super HDTVsignal is assigned to six times domains. If a NTSC compressed signal is6 Mbps, this assignment can carry as nine times higher as 54 Mbps of D₃data. Accordingly, super HDTV data of higher picture quality can betransmitted.

The foregoing data assignment makes the use of one of two, horizontaland vertical, polarization planes of a transmission wave. When both thehorizontal and vertical polarization planes are used, the frequencyutilization will be doubled. This will be explained below.

FIG. 49 shows a data assignment in which D_(V1) and D_(H1) are avertical and a horizontal polarization signal of the first data streamrespectively, D_(V2) and D_(H2) are a vertical and a horizontalpolarization signal of the second data stream respectively, and D_(V3)and D_(H3) are a vertical and a horizontal polarization signal of thethird data stream respectively. The vertical, polarization signal D_(V1)of the first data stream carries a low frequency band or NTSC TV dataand the horizontal polarization signal D_(H1) carries a high frequencyband or HDTV data. When the first receiver 23 is equipped with avertical polarization antenna, it can reproduce only the NTSC signal.When the first receiver 23 is equipped with an antenna for bothhorizontally and vertically polarized waves, it can reproduce the HDTVsignal through summing L1 and M1. More specifically, the first receiver23 can provide compatibility between NTSC and HDTV with the use of aparticular type antenna.

FIG. 50 illustrates a TDMA method in which each data burst 721 isaccompanied at front a sync data 731 and a card data 741. Also, a framesync data 720 is provided at the front of a frame. Like channels areassigned to like time slots. For example, a first time slot 750 carriesNTSC, HDTV, and super HDTV data of the first channel simultaneously. Thesix time slots 750, 750 a, 750 b, 750 c, 750 d, 750 e are arrangedindependent from each other. Hence, each station can offer NTSC, HDTV,and/or supper HDTV services independently of the other stations throughselecting a particular channel of the time slots. Also, the firstreceiver 23 can reproduce an NTSC signal when equipped with a horizontalpolarization antenna and both NTSC and HDTV signals when equipped with acompatible polarization antenna. In this respect, the second receiver 33can reproduce a super HDTV at lower resolution while the third receiver43 can reproduce a full super HDTV signal. According to the thirdembodiment, a compatible signal transmission system will be constructed.It is understood that the data assignment is not limited to the burstmode TDMA method shown in FIG. 50 and another method such as timedivision multiplexing of continuous signals as shown in FIG. 49 will beemployed with equal success. Also, a data assignment shown in FIG. 51will permit a HDTV signal to be reproduced at high resolution.

As set forth above, the compatible digital TV signal transmission systemof the third embodiment can offer three, super HDTV, HDTV, andconventional NTSC, TV broadcast services simultaneously. In addition, avideo signal intercepted by a commercial station or cinema can beelectronized.

The modified QAM of the embodiments is now termed as SRQAM and its errorrate will be examined.

First, the error rate in 16 SRQAM will be calculated. FIG. 99 shows avector diagram of 16 SRQAM signal points. As apparent from the firstquadrant, the 16 signal points of standard 16 QAM including 83 a, 83 b,84 a, 83 a are allocated at equal intervals of 2δ.

The signal point 83 a is spaced δ from both the I-axis and the Q-axis ofthe coordinate. It is now assumed that n is a shift value of the 16SRQAM. In 16 SRQAM, the signal point 83 a of 16 QAM is shifted to asignal point 83 where the distance from each axis is nδ. The shift valuen is thus expressed as:0≦n≦3.

The other signal points 84 a and 86 a are also shifted to two points 84and 86 respectively.

If the error rate of the first data stream is Pe₁, it is obtained from:$\begin{matrix}{{{{Pe}\quad 1} - 16} = {{\frac{1}{4}\text{erfc}( \frac{n\quad\delta}{\sqrt{2\quad\sigma}} )} + {\text{erfc}( \frac{3\delta}{\sqrt{2\sigma}} )}}} \\{= {\frac{1}{8}\text{erfc}( \frac{n\sqrt{\rho}}{\sqrt{9 + n^{2}}} )}}\end{matrix}$Also, the error rate Pe₂ of the second data stream is obtained from:$\begin{matrix}{{{{Pe}\quad 2} - 16} = {\frac{1}{2}\text{erfc}( \frac{\frac{3 \cdot n}{2}\quad\delta}{\sqrt{2\quad\sigma}} )}} \\{= {\frac{1}{4}\text{erfc}( {\frac{3 - n}{2\sqrt{9 + n^{2}}}\sqrt{\rho}} )}}\end{matrix}$

The error rate of 36 or 32 SRQAM will be calculated. FIG. 100 is avector diagram of a 36 SRQAM signal in which the distance between anytwo 36 QAM signal points is 2δ.

The signal point 83 a of 36 QAM is spaced δ from each axis of thecoordinate. It is now assumed that n is a shift value of the 16 SRQAM.In 36 SRQAM, the signal point 83 a is shifted to a signal point 83 wherethe distance from each axis is nδ. Similarly, the nine 36 QAM signalpoints in the first quadrant are shifted to points 83, 84, 85, 86, 97,98, 99, 100, 101 respectively. If a signal point group 90 comprising thenine signal points is regarded as a single signal point, the error ratePe₁ in reproduction of only the first data stream D₁ with a modified 4PSK receiver and the error rate Pe₂ in reproduction of the second datastream D₂ after discriminating the nine signal points of the group 90from each other, are obtained respectively from: $\begin{matrix}{{{{Pe}\quad 1} - 32} = {\frac{1}{6}{{erfc}( \frac{n\quad\delta}{\sqrt{2\sigma}} )}}} \\{= {\frac{1}{6}{{erfc}( {\sqrt{\frac{6\quad\rho}{5}} \times \frac{n}{\sqrt{n^{2} + {2n} + 25}}} )}}}\end{matrix}$ $\begin{matrix}{{{{Pe}\quad 2} - 32} = {\frac{2}{3}{{erfc}( {\frac{5 - n}{4\sqrt{2}}\frac{\delta}{\rho}} )}}} \\{= {\frac{2}{3}{{erfc}( {\sqrt{\frac{3\quad\rho}{40}} \times \frac{5 - n}{\sqrt{n^{2} + {2n} + 25}}} )}}}\end{matrix}$

FIG. 101 shows the relation between error rate Pe and C/N rate intransmission in which the curve 900 represents a conventional or notmodified 32 QAM signal. The straight line 905 represents a signal having10^(−1.5) of the error rate. The curve 901 a represents a D₁ level 32SRQAM signal of the present invention at the shift rate n of 1.5. Asshown, the C/N rate of the 32 SRQAM signal is 5 dB lower at the errorrate of 10^(−1.5) than that of the conventional 32 QAM. This means thatthe present invention allows a D₁ signal to be reproduced at a givenerror rate when its C/N rate is relatively low.

The curve 902 a represents a D₂ level SRQAM signal at n=1.5 which can bereproduced at the error rate of 10⁻¹⁵ only when its C/N rate is 2.5 dBhigher than that of the conventional 32 QAM of the curve 900. Also, thecurves 901 b and 902 b represent D₁ and D₂ SRQAM signals at n=2.0respectively. The curves 902 c represents a D₂ SRQAM signal at n=2.5. Itis apparent that the C/N rate of the SRQAM signal at the error rate of10^(−1.5) is 5 dB, 8 dB, and 10 dB higher at n=1.5, 2.0, and 2.5respectively in the D₁ level and 2.5 dB lower in the D₂ level than thatof a common 32 QAM signal.

Shown in FIG. 103 is the C/N rate of the first and second data streamsD₁, D₂ of a 32 SRQAM signal which is needed for maintaining a constanterror rate against variation of the shift n. As apparent, when the shiftn is more than 0.8, there is developed a clear difference between twoC/N rates of their respective D₁ and D₂ levels so that the multi-levelsignal, namely first and second data, transmission can be implementedsuccessfully. In brief, n>0.85 is essential for multi-level datatransmission of the 32 SRQAM signal of the present invention.

FIG. 102 shows the relation between the C/N rate and the error rate for16 SRQAM signals. The curve 900 represents a common 16 QAM signal. Thecurves 901 a, 901 b, 901 c and D₁ level or first data stream 16 SRQAMsignals at n=1.2, 1.5, and 1.8 respectively. The curves 902 a, 902 b,902 c are D₂ level or second data stream 16 SRQAM signals at n=1.2, 1.5,and 1.8 respectively.

The C/N rate of the first and second data streams D₁, D₂ of a 16 SRQAMsignal is shown in FIG. 104, which is needed for maintaining a constanterror rate against variation of the shift n. As apparent, when the shiftn is more than 0.9 (n>0.9), the multi-level data transmission of the 16SRQAM signal will be executed.

One example of propagation of SRQAM signals of the present inventionwill now be described for use with a digital TV terrestrial broadcastservice. FIG. 105 shows the relation between the signal level and thedistance between a transmitter antenna and a receiver antenna in theterrestrial broad cast service. The curve 911 represents a transmittedsignal from the transmitter antenna of 1250 feet high. It is assumedthat the error rate essential for reproduction of an applicable digitalTV signal is 10^(−1.5). The hatching area 912 represents a noiseinterruption. The point 910 represents a signal reception limit of aconventional 32 QAM signal at C/N=15 dB where the distance L is 60 milesand a digital HDTV signal can be intercepted at minimum.

The C/N rate varies 5 dB under a worse receiving condition such as badweather. If a change in the relevant condition, e.g. weather, attenuatesthe C/N rate, the interception of an HDTV signal will hardly be ensured.Also, geographical conditions largely affect the propagation of signalsand a decrease of about 10 dB at least will be unavoidable. Hence,successful signal interception within 60 miles will never be guaranteedand above all, a digital signal will be propagated harder than ananalogue signal. It would be understood that the service area of aconventional digital TV broadcast service is less dependable.

In case of the 32 SRQAM signal of the present invention, three-levelsignal transmission system is constituted as shown in FIGS. 133 and 137.This permits a low resolution NTSC signal of MPEG level to be carried onthe 1-1 data stream D₁₋₁, a medium resolution TV data of e.g. NTSCsystem to be carried on the 1-2 data stream D₁₋₂, and a high frequencycomponent of HDTV data to be carried on the second data stream D₂.Accordingly, the service area of the 1-2 data stream of the SRQAM signalis increased to a 70 mile point 910 a while of the second data streamremains within a 55 mile point 910 b, as shown in FIG. 105. FIG. 106illustrates a computer simulation result of the service area of the 32SRQAM signal of the present invention, which is similar to FIG. 53 butexplains in more detail. As shown, the regions 708, 703 c, 703 a, 703 b,712 represent a conventional 32 QAM receivable area, a 1-1 data levelD₁₋₁ receivable area, a 1-2 data level D₁₋₂ receivable area, a seconddata level D₂ receivable area, and a service area of a neighbor analogueTV station respectively. The conventional 32 QAM signal data used inthis drawing is based on a conventionally disclosed one.

For common 32 QAM signal, the 60-mile-radius service area can beestablished theoretically. The signal level will however be attenuatedby geographical or weather conditions and particularly, considerablydeclined at near the limit of the service area.

If the low frequency band TV component of MPEG1 grade is carried on the1-1 level D₁₋₁ data and the medium frequency band TV component of NTSCgrade on the 1-2 level D₁₋₂ data and high frequency band TV component ofHDTV on the second level D₂ data, the service area of the 32 SRQAMsignal of the present invention is increased by 10 miles in radius forreception of an EDTV signal of medium resolution grade and 18 miles forreception of an LDTV signal of low resolution grade although decreasedby 5 miles for reception of an HDTV signal of high resolution grade, asshown in FIG. 106. FIG. 107 shows a service area in case of a shiftfactor n or s=1.8. FIG. 135 shows the service area of FIG. 107 in termsof area.

More particularly, the medium resolution component of a digital TVbroadcast signal of the SRQAM mode of the preset invention cansuccessfully be intercepted in an unfavorable service region or shadowarea where a conventional medium frequency band TV signal is hardlypropagated and attenuated due to obstacles. Within at least thepredetermined service area, the NTSC TV signal of the SRQAM mode can beintercepted by any traditional TV receiver. As the shadow or signalattenuating area developed by building structures and other obstacles orby interference of a neighbor analogue TV signal or produced in a lowland is decreased to a minimum, TV viewers or subscribers will beincreased in number.

Also, the HDTV service can be appreciated by only a few viewers whoafford to have a set of high cost HDTV receiver and display, accordingto the conventional system. The system of the present invention allows atraditional NTSC, PAL, or SECAM receiver to intercept a mediumresolution component of the digital HDTV signal with the use of anadditional digital tuner. A majority of TV viewers can hence enjoy theservice at less cost and will be increased in number. This willencourage the TV broadcast business and create an extra social benefit.

Furthermore, the signal receivable area for medium resolution or NTSC TVservice according to the present invention is increased about 36% atn=2.5, as compared with the conventional system. As the service areathus the number of TV viewers is increased, the TV broadcast businessenjoys an increasing profit. This reduces a risk in the development of anew digital TV business which will thus be encouraged to put intopractice.

FIG. 107 shows the service area of a 32 SRQAM signal of the presentinvention in which the same effect will be ensured at n=1.8. Two serviceareas 703 a, 703 b of D₁ and D₂ signals respectively can be determinedin extension for optimum signal propagation by varying the shift nconsidering a profile of HDTV and NTSC receiver distribution orgeographical features. Accordingly, TV viewers will satisfy the serviceand a supplier station will enjoy a maximum of viewers.

This advantage is given when:n>1.0Hence, if the 32 SRQAM signal is selected, the shift n is determined by:1<n<5Also, if the 16 SRQAM signal is employed, n is determined by:1<n<3In the SRQAM mode signal terrestrial broadcast service in which thefirst and second data levels are created by shifting correspondingsignal points as shown in FIGS. 99 and 100, the advantage of the presentinvention will be given when the shift n in a 16, 32, or 64 SRQAM signalis more than 1.0.

In the above embodiments, the low and high frequency band components ofa video signal are transmitted as the first and second data streams.However, the transmitted signal may be an audio signal. In this case,low frequency or low resolution components of an audio signal may betransmitted as the first data stream, and high frequency or highresolution components of the audio signal may be transmitted as thesecond data stream. Accordingly, it is possible to receive high C/Nportion in high sound quality, and low C/N portion in low sound quality.This can be utilized in PCM broadcast, radio, portable telephone and thelike. In this case, the broadcasting area or communication distance canbe expanded as compared with the conventional systems.

Furthermore, the third embodiment can incorporate a time divisionmultiplexing (TDM) system as shown in FIG. 133. Utilization of the TDMmakes it possible to increase the number of subchannels. An ECC encoder743 a and ECC encoder 743 b, provided in two subchannels, differentiateECC code gains so as to make a difference between thresholds of thesetwo subchannels. Whereby, an increase of channel number of themulti-level signal transmission can be realized. In this case, it isalso possible to provide two Trellis encoders 743 a, 743 b as shown inFIG. 137 and differentiate their code gains. The explanation of thisblock diagram is substantially identical to that of later describedblock diagram of FIG. 131 which shows the sixth embodiment of thepresent invention and, therefore, will not be described here.

In a simulation of FIG. 106, there is provided 5 dB difference of acoding gain between 1-1 subchannel D₁₋₁ and 1-2 subchannel D₁₋₂.

An SRQAM is the system applying a C-CDM (Constellation-Code DivisionMultiplex) of the present invention to a rectangle-QAM. A C-CDM, whichis a multiplexing method independent of TDM or FDM, can obtainsubchannels by dividing a constellation-code corresponding to a code. Anincrease of the number of codes will bring an expansion of transmissioncapacity, which is not attained by TDM or FDM alone, while maintainingalmost perfect compatibility with conventional communication apparatus.Thus C-CDM can bring excellent effects.

Although above embodiment combines the C-CDM and the TDM, it is alsopossible to combine the C-CDM with the FDM (Frequency DivisionMultiplex) to obtain similar modulation effect of threshold values. Sucha system can be used for a TV broadcasting, and FIG. 108 shows afrequency distribution of a TV signal. A spectrum 725 represents afrequency distribution of a conventional analogue, e.g. NTSC,broadcasting signal. The largest signal is a video carrier 722. A colorcarrier 723 and a sound carrier 724 are not so large. There is known amethod of using an FDM for dividing a digital broadcasting signal intotwo frequencies. In this case, a carrier is divided into a first carrier726 and a second carrier 727 to transmit a first 720 and a second signal721 respectively. An interference can be lowered by placing first andsecond carriers 726, 727 sufficiently far from the video carrier 722.The first signal 720 serves to transmit a low resolution TV signal at alarge output level, while the second signal 721 serves to transmit ahigh resolution TV signal at a small output level. Consequently, themulti-level signal transmission making use of an FDM can be realizedwithout being bothered by obstruction.

FIG. 134 shows an example of a conventional method using a 32 QAMsystem. As the subchannel A has a larger output than the subchannel B, athreshold value for the subchannel A, i.e. a threshold 1, can be setsmall 4˜5 dB than a threshold value for the subchannel B, i.e. athreshold 2. Accordingly, a two-level broadcasting having 4˜5 dBthreshold difference can be realized. In this case, however, a largereduction of signal reception amount will occur if the receiving signallevel decreases below the threshold 2. Because the second signal 721 a,having a large information amount as shaded in the drawing, cannot bereceived in such a case and only the first signal 720 a, having a smallinformation amount, is received. Consequently, a picture quality broughtby the second level will be extremely worse.

However, the present invention resolves this problem. According to thepresent invention, the first signal 720 is given by 32 SRQAM mode whichis obtained through C-CDM modulation so that the subchannel A is dividedinto two subchannels 1 of A and 2 of A. The newly added subchannel 1 ofA, having a lowest threshold value, carries a low resolution component.The second signal 721 is also given by 32 SRQAM mode, and a thresholdvalue for the subchannel 1 of B is equalized with the threshold 2.

With this arrangement, the region in which a transmitted signal is notreceived when the signal level decreases below the threshold 2 isreduced to a shaded portion of the second signal 721 a in FIG. 108. Asthe subchannel 1 of B and the subchannel A are both receivable, thetransmission amount is not so much reduced in total. Accordingly, abetter picture quality is reproduced even in the second level at thesignal level of the threshold 2.

By transmitting a normal resolution component in one subchannel, itbecomes possible to increase the number of multiple level and expand alow resolution service area. This low-threshold subchannel is utilizedfor transmitting important information such as sound information, syncinformation, headers of respective data, because these informationcarried on this low-threshold subchannel can be surely received. Thusstable reception is feasible. If a subchannel is newly added in thesecond signal 721 in the same manner, the level number of multi-leveltransmission can be increased in the service area. In the case where anHDTV signal has 1050 scanning lines, an new service area equivalent to775 lines can be provided in addition to 525 lines.

Accordingly, the combination of the FDM and the C-CDM realizes anincrease of service area. Although above embodiment divides a subchannelinto two, it is needless to say it will also be preferable to divide itinto three or more.

Next, a method of avoiding obstruction by combining the TDM and theC-CDM will be explained. As shown in FIG. 109, an analogue TV signalincludes a horizontal retrace line portion 732 and a video signalportion 731. This method utilizes a low signal level of the horizontalretrace line portion 732 and non-display of obstruction on a pictureplane during this period. By synchronizing a digital TV signal with ananalogue TV signal, horizontal retrace line sync slots 733, 733 a of thehorizontal retrace line portion 732 can be used for transmission of animportant, e.g. a sync, signal or numerous data at a high output level.Thus, it becomes possible to increase data amount or output levelwithout increasing obstruction. The similar effect will be expected evenif vertical retrace line sync slots 737, 737 a are providedsynchronously with vertical retrace line portions 735, 735 a.

FIG. 110 shows a principle of the C-CDM. Furthermore,

FIG. 111 shows a code assignment of the C-CDM equivalent to an expanded16 QAM. FIG. 112 shows a code assignment of the C-CDM equivalent to anexpanded 32 QAM. As shown in FIGS. 110 and 111, a 256 QAM signal isdivided into four, 740 a, 740 b, 740 c, 740 d, levels which have 4, 16,64, 256 segments, respectively. A signal code word 742 d of 256 QAM onthe fourth level 740 d is “11111111” of 8 bit. This is split into fourcode words 741 a, 741 b, 741 c, and 741 d of 2-bit - - - i.e. “11”,“11”, “11”, “11”, which are then allocated on signal point regions 742a, 742 b, 742 c, 742 d of first, second, third, fourth levels 740 a, 740b, 740 c, 740 d, respectively. As a result, subchannels 1, 2, 3, 4 of 2bit are created. This is termed as C-CDM (Constellation-Code DivisionMultiplex). FIG. 111 shows a detailed code assignment of the C-CDMequivalent to expanded 16 QAM, and FIG. 112 shows a detailed codeassignment of the C-CDM equivalent to expanded 32 QAM. As the C-CDM isan independent multiplexing method, it can be combined with theconventional FDM (Frequency Division Multiplex) or TDM (Time DivisionMultiplex) to further increase the number of subchannels. In thismanner, the C-CDM method realizes a novel multiplexing system. Althoughthe C-CDM is explained by using rectangle QAM, other modulation systemhaving signal points, e.g. QAM, PSK, ASK, and even FSK if frequencyregions are regarded as signal points, can be also used for thismultiplexing in the same manner.

For example, the error rate of the subchannel 1 of 8PS-APSK, explainedin the embodiment 1 with reference to FIG. 139, will be expressed asfollow:${{{Pe}\quad 1} - 8} = {{\frac{1}{4}\text{erfc}( \frac{\delta}{\sqrt{2}\delta} )} + {\frac{1}{4}\text{erfc}( \frac{( {S_{1} + 1} )\delta}{\sqrt{2}\sigma} )}}$

The error rate of the subchannel 2 is expressed as follows:${{{Pe}\quad 2} - 8} = {\frac{1}{2}\text{erfc}( \frac{S_{1}\delta}{2\sigma} )}$

Furthermore, the error rate of the subchannel 1 of 16-PS-APSK (PS type),explained with reference to FIG. 142, will be expressed as follow:${{{Pe}\quad 1} - 16} = {{\frac{1}{8}\text{erfc}( \frac{\delta}{\sqrt{2}\sigma} )} + {\frac{1}{8}\text{erfc}( \frac{( {S_{2} + 1} )\delta}{\sqrt{2}\sigma} )} + {\frac{1}{8}{{erfc}( \frac{( {S_{1} + 1} )\delta}{\sqrt{2}\sigma} )}} + {\frac{1}{8}{{erfc}( \frac{( {S_{1} + S_{2} + 1} )\delta}{\sqrt{2}\sigma} )}}}$

The error rate of the subchannel 2 is expressed as follows:${{{Pe}\quad 2} - 16} = {{\frac{1}{4}\text{erfc}( \frac{S_{1}\delta}{2\sigma} )} + {\frac{1}{8}\text{erfc}( \frac{( {S_{1} - S_{2}} )\delta}{2\sigma} )} + {\frac{1}{8}\text{erfc}( \frac{( {S_{1} + S_{2}} )\delta}{2\sigma} )}}$

The error rate of the subchannel 3 is expressed as follows:${{{Pe}\quad 3} - 10} = {\frac{1}{2}\text{erfc}( \frac{S_{2}\delta}{2\sigma} )}$

Embodiment 4

A fourth embodiment of the present invention will be described referringto the relevant drawings.

FIG. 37 illustrates the entire arrangement of a signal transmissionsystem of the fourth embodiment, which is arranged for terrestrialservice and similar in both construction and action to that of the thirdembodiment shown in FIG. 29. The difference is that the transmitterantenna 6 is replaced with a terrestrial antenna 6 a and the receiverantennas 22, 23, 24 are replaced with also three terrestrial antennas 22a, 23 a, 24 a. The action of the system is identical to that of thethird embodiment and will no more be explained. The terrestrialbroadcast service unlike a satellite service depends much on thedistance between the transmitter antenna 6 a to the receiver antennas 22a, 32 a, 42 a. If a receiver is located far from the transmitter, thelevel of a received signal is low. Particularly, a common multi-levelQAM signal can hardly be demodulated by the receiver which thusreproduces no TV program.

The signal transmission system of the present invention allows the firstreceiver 23 equipped with the antenna 22 a, which is located at a fardistance as shown in FIG. 37, to intercept a modified 16 or 64 QAMsignal and demodulate at 4 PSK mode the first data stream or D₁component of the received signal to an NTSC video signal so that a TVprogram picture of medium resolution can be displayed even if the levelof the received signal is relatively low.

Also, the second receiver 33 with the antenna 32 a is located at amedium distance from the antenna 6 a and can thus intercept anddemodulate both the first and second data streams or D₁ and D₂components of the modified 16 or 64 QAM signal to an HDTV video signalwhich in turn produces an HDTV program picture.

The third receiver 43 with the antenna 42 a is located at a neardistance and can intercept and demodulate the first, second, and thirddata streams or D₁, D₂, and D₃ components of the modified 16 or 64 QAMsignal to a super HDTV video signal which in turn produces a super HDTVpicture in quality to a common movie picture.

The assignment of frequencies is determined by the same manner as of thetime division multiplexing shown in FIGS. 34, 35, and 36. Like FIG. 34,when the frequencies are assigned t first to sixth channels, L1 of theD₁ component carries an NTSC data of the first channel, M1 of the D2component carries an HDTV difference data of the first channel, and H1of the D₃ component carries a super HDTV difference data of the firstchannel. Accordingly, NTSC, HDTV, and super HDTV data all can be carriedon the same channel. If D₂ and D₃ of the other channels are utilized asshown in FIGS. 35 and 36, more data of HDTV and super HDTV respectivelycan be transmitted for higher resolution display.

As understood, the system allows three different but compatible digitalTV signals to be carried on a single channel or using D₂ and D₃ regionsof other channels. Also, the medium resolution TV picture data of eachchannel can be intercepted in a wider service area according to thepresent invention.

A variety of terrestrial digital TV broadcast systems employing a 16 QAMHDTV signal of 6 MHz bandwidth have been proposed. Those are however notcompatible with the existing NTSC system and thus, have to be associatedwith a simulcast technique for transmitting NTSC signals of the sameprogram on another channel. Also, such a common 16 QAM signal limits aservice area. The terrestrial service system of the present inventionallows a receiver located at a relatively far distance to interceptsuccessfully a medium resolution TV signal with no use of an additionaldevice nor an extra channel.

FIG. 52, shows an interference region of the service area 702 of aconventional terrestrial digital HDTV broadcast station 701. As shown,the service area 702 of the conventional HDTV station 701 is intersectedwith the service area 712 of a neighbor analogue TV station 711. At theintersecting region 713, an HDTV signal is attenuated by signalinterference from the analogue TV station 711 and will thus beintercepted with less consistency.

FIG. 53 shows an interference region associated with the multi-levelsignal transmission system of the present invention. The system is lowin the energy utilization as compared with a conventional system and itsservice area 703 for HDTV signal propagation is smaller than the area702 of the conventional system. In contrary, the service area 704 fordigital NTSC or medium resolution TV signal propagation is larger thanthe conventional area 702. The level of signal interference from adigital TV station 701 of the system to a neighbor analogue TV station711 is equivalent to that from a conventional digital TV station, suchas shown in FIG. 52.

In the service area of the digital TV station 701, there are threeinterference regions developed by signal interference from the analogueTV station 711. Both HDTV and NTSC signals can hardly be intercepted inthe first region 705. Although fairly interfered, an NTSC signal may beintercepted at an equal level in the second region 706 denoted by theleft down hatching. The NTSC signal is carried on the first data streamwhich can be reproduced at a relatively low C/N rate and will thus beminimum affected when the C/N rate is declined by signal interferencefrom the analogue TV station 711.

At the third region 707 denoted by the right down hatching, an HDTVsignal can also be intercepted when signal interference is absent whilethe NTSC signal can constantly be intercepted at a low level.

Accordingly, the overall signal receivable area of the system will beincreased although the service area of HDTV signals becomes a little bitsmaller than that of the conventional system. Also, at the signalattenuating regions produced by interference from a neighbor analogue TVstation, NTSC level signals of an HDTV program can successfully beintercepted as compared with the conventional system where no HDTVprogram is viewed in the same area. The system of the present inventionmuch reduces the size of signal attenuating area and when increases theenergy of signal transmission at a transmitter or transponder station,can extend the HDTV signal service area to an equal size to theconventional system. Also, NTSC level signals of a TV program can beintercepted more or less in a far distance area where no service isgiven by the conventional system or a signal interference area caused byan adjacent analogue TV station.

Although the embodiment employs a two-level signal transmission method,a three-level method such as shown in FIG. 78 will be used with equalsuccess. If an HDTV signal is divided into three picture levels-HDTV,NTC, and low resolution NTSC, the service area shown in FIG. 53 will beincreased from two levels to three levels where the signal propagationis extended radially and outwardly. Also, low resolution NTSC signalscan be received at an acceptable level at the first signal interferenceregion 705 where NTSC signals are hardly be intercepted in the two-levelsystem. As understood, the signal interference is also involved from adigital TV station to an analogue TV station.

The description will now be continued, provided that no digital TVstation should cause a signal interference to any neighbor analogue TVstation. According to a novel system under consideration in U.S.A.,no-use channels of the existing service channels are utilized for HDTVand thus, digital signals must not interfere with analogue signals. Forthe purpose, the transmitting level of a digital signal has to bedecreased lower than that shown in FIG. 53. If the digital signal is ofconventional 16 QAM or 4 PSK mode, its HDTV service area 708 becomesdecreased as the signal interference region 713 denoted by the crosshatching is fairly large as shown in FIG. 54. This results in a lessnumber of viewers and sponsors, whereby such a digital system will havemuch difficulty to operate for profitable business.

FIG. 55 shows a similar result according to the system of the presentinvention. As apparent, the HDTV signal receivable 703 is a little bitsmaller than the equal area 708 of the conventional system. However, thelower resolution or NTSC TV signal receivable area 704 will be increasedas compared with the conventional system. The hatching area represents aregion where the NTSC level signal of a program can be received whilethe HDTV signal of the same is hardly intercepted. At the firstinterference region 705, both HDTV and NTSC signals cannot beintercepted due to signal interference from an analogue station 711.

When the level of signals is equal, the multi-level transmission systemof the present invention provides a smaller HDTV service area and agreater NTSC service area for interception of an HDTV program at an NTSCsignal level. Accordingly, the overall service area of each station isincreased and more viewers can enjoy its TV broadcasting service.Furthermore, HDTV/NTSC compatible TV business can be operated witheconomical advantages and consistency. It is also intended that thelevel of a transmitting signal is increased when the control on avertingsignal interference to neighbor analogue TV stations is lessenedcorresponding to a sharp increase in the number of home-use digitalreceivers. Hence, the service area of HDTV signals will be increased andin this respect, the two different regions for interception of HDTV/NTSCand NTSC digital TV signal levels respectively, shown in FIG. 55, can beadjusted in proportion by varying the signal point distance in the firstand/or second data stream. As the first data stream carries informationabout the signal point distance, a multi-level signal can be receivedwith more certainty.

FIG. 56 illustrates signal interference between two digital TV stationsin which a neighbor TV station 701 a also provides a digital TVbroadcast service, as compared with an analogue station in FIG. 52.Since the level of a transmitting signal becomes high, the HDTV serviceor high resolution TV signal receivable area 703 in increased to anextension equal to the service area 702 of an analogue TV system.

At the intersecting region 714 between two service areas of theirrespective stations, the received signal can be reproduced not to anHDTV level picture with the use of a common directional antenna due tosignal interference but to an NTSC level picture with a particulardirectional antenna directed towards a desired TV station. If a highlydirectional antenna is used, the received signal from a target stationwill be reproduced to an HDTV picture. The low resolution signalreceivable area 704 is increased larger than the analogue TV systemservice area 702 and a couple of intersecting regions 715, 716 developedby the two low resolution signal receivable areas 704 and 704 a of theirrespective digital TV stations 701 and 701 a permit the received signalfrom antenna directed one of the two stations to be reproduced to anNTSC level picture.

The HDTV service area of the multi-level signal transmission system ofthe present invention itself will be much increased when applicablesignal restriction rules are withdrawn in a coming digital TV broadcastservice maturity time.

At the time, the system of the present invention also provides as a wideHDTV signal receivable area as of the conventional system andparticularly, allows its transmitting signal to be reproduced at an NTSClevel in a further distance or intersecting areas where TV signals ofthe conventional system are hardly intercepted. Accordingly, signalattenuating or shadow regions in the service area will be minimized.

Embodiment 5

A first embodiment of the present invention resides in amplitudemodulation or ASK procedure. FIG. 57 illustrates the assignment ofsignal points of a 4-level ASK signal according to the fifth embodiment,in which four signal points are denoted by 721, 722, 723, and 724. Thefour-level transmission permits a 2-bit data to be transmitted in everycycle period. It is assumed that the four signal points 721, 722, 723,724 represent two-bit patterns 00, 01, 10, 11 respectively.

For ease of four-level signal transmission of the embodiment, the twosignal points 721, 722 are designated as a first signal point group 725and the other two 723, 724 are designated as a second signal point group726. The distance between the two signal point groups 725 and 726 isthen determined wider than that between any two adjacent signal points.More specifically, the distance L₀ between the two signals 722 and 723is arranged wider than the distance L between the two adjacent points721 and 722 or 723 and 724.

This is expressed as:L₀>LHence, the multi-level signal transmission system of the embodiment isbased on L₀>L. The embodiment is however not limited to L₀>L and L=L₀will be employed temporarily or permanently depending on therequirements of design, condition, and setting.

The two signal point groups are assigned one-bit patterns of the firstdata stream D₁, as shown in FIG. 59(a). More particularly, a bit 0 ofbinary system is assigned to the first signal point group 725 andanother bit 1 to the second signal point group 726. Then, a one-bitpattern of the second data stream D₂ is assigned to each signal point.For example, the two signal points 721, 723 are assigned D₂=0 and theother two signal points 722 and 724 are assigned D₂=1. Those are thusexpressed by two bits per symbol.

The multi-level signal transmission of the present invention can beimplemented in an ASK mode with the use of the foregoing signal pointassignment. The system of the present invention works in the same manneras of a conventional equal signal point distance technique when thesignal to noise ratio or C/N rate is high. If the C/N rate becomes lowand no data can be reproduced by the conventional technique, the presentsystem ensures reproduction of the first data stream D₁ but not thesecond data stream D₂. In more detail, the state at a low C/N is shownin FIG. 60. The signal points transmitted are displaced by a Gaussiandistribution to ranges 721 a, 722 a, 723 a, 724 a respectively at thereceiver side due to noise and transmission distortion. Therefore, thedistinction between the two signals 721 and 722 or 723 and 724 willhardly be executed. In other words, the error rate in the second datastream D₂ will be increased. As apparent from FIG. 60, the two signalpoints 721, 722 are easily distinguished from the other two signalpoints 723, 724. The distinction between the two signal point groups 725and 726 can thus be carried out with ease. As the result, the first datastream D₁ will be reproduced at a low error rate.

Accordingly, the two different level data D₁ and D₂ can be transmittedsimultaneously. More particularly, both the first and second datastreams D₁ and D₂ of a given signal transmitted through the multi-leveltransmission system can be reproduced at the area where the C/N rate ishigh and the first data stream D₁ only can be reproduced in the areawhere the C/N rate is low.

FIG. 61 is a block diagram of a transmitter 741 in which an input unit742 comprises a first data stream input 743 and a second data streaminput 744. A carrier wave from a carrier generator 64 is amplitudemodulated by a multiplier 746 using an input signal fed across aprocessor 745 from the input unit 743. The modulated signal is then bandlimited by a filter 747 to an ASK signal of e.g. VSB mode which is thendelivered from an output unit 748.

The waveform of the ASK signal after filtering will now be examined.FIG. 62(a) shows a frequency spectrum of the ASK modulated signal inwhich two sidebands are provided on both sides of the carrier frequencyband. One of the two sidebands is eliminated with the filter 474 toproduce a signal 749 which contains a carrier component as shown in FIG.62(b). The signal 749 is a VSB signal and if the modulation frequencyband is f₀, will be transmitted in a frequency band of about f₀/2.Hence, the frequency utilization becomes high. Using VSB modetransmission, the ASK signal of two bit per symbol shown in FIG. 60 canthus carry in the frequency band an amount of data equal to that of 16QAM mode at four bits per symbol.

FIG. 63 is a block diagram of a receiver 751 in which an input signalintercepted by a terrestrial antenna 32 a is transferred through aninput unit 752 to a mixer 753 where it is mixed with a signal from avariable oscillator 754 controlled by channel selection to a lowermedium frequency signal. The signal from the mixer 753 is then detectedby a detector 755 and filtered by an LPF 756 to a baseband signal whichis transferred to a discriminating/reproduction circuit 757. Thediscrimination/reproduction circuit 757 reproduces two, first D₁ andsecond D₂, data streams from the baseband signal and transmit themfurther through a first 758 and a second data stream output 759respectively.

The transmission of a TV signal using such a transmitter and a receiverwill be explained. FIG. 64 is a block diagram of a video signaltransmitter 774 in which a high resolution TV signal, e.g. an HDTVsignal, is fed through an input unit 403 to a divider circuit 404 of afirst video encoder 401 where it is divided into four high/low frequencyTV signal components denoted by e.g. H_(L)V_(L), H_(L)V_(H), H_(H)V_(L),and H_(H)V_(H). This action is identical to that of the third embodimentpreviously described referring to FIG. 30 and will no more be explainedin detail. The four separate TV signals are encoded respectively by acompressor 405 using a known DPCMDCT variable length code encodingtechnique which is commonly used e.g. in MPEG. Meanwhile, the motioncompensation of the signal is carried out at the input unit 403. Thecompressed signals are summed by a summer 771 to two, first and second,data streams D₁, D₂. The low frequency video signal component orH_(L)V_(L) signal is contained in the first data stream D₁. The two datastream signals D₁, D₂ are then transferred to a first 743 and a seconddata stream input 744 of a transmitter unit 741 where they are amplitudemodulated and summed to an ASK signal of e.g. VSB mode which ispropagated from a terrestrial antenna for broadcast service.

FIG. 65 is a block diagram of a TV receiver for such a digital TVbroadcast system. A digital TV signal intercepted by a terrestrialantenna 32 a is fed to an input 752 of a receiver 781. The signal isthen transferred to a detection/demodulation circuit 760 where a desiredchannel signal is selected and demodulated to two, first and second,data streams D₁, D₂ which are then fed to a first 758 and a second datastream output 759 respectively. The action in the receiver unit 751 issimilar to that described previously and will no more be explained indetail. The two data streams D₁, D₂ are sent to a divider unit 776 inwhich D₁ is divided by a divider 777 into two components; one orcompressed H_(L)V_(L) is transferred to a first input 521 of a secondvideo decoder 422 and the other is fed to a summer 778 where it issummed with D₂ prior to transfer to a second input 531 of the secondvideo decoder 422. Compressed H_(L)V_(L) is then sent from the firstinput 521 to a first expander 523 where it is expanded to H_(L)V_(L) ofthe original length which is then transferred to a video mixer 548 andan aspect ratio changing circuit 779. When the input TV signal is anHDTV signal, H_(L)V_(L) represents a wide-screen NTSC signal. When thesame is an NTSC signal, H_(L)V_(L) represents a lower resolution videosignal, e.g. MPEG1, that an NTSC level.

The input TV signal of the embodiment is an HDTV signal and H_(L)V_(L)becomes a wide-screen NTSC signal. If the aspect ratio of an availabledisplay is 16:9, H_(L)V_(L) is directly delivered through an output unitas a 16:9 video output 426. If the display has an aspect ratio of 4:3,H_(L)V_(L) is shifted by the aspect ratio changing circuit 779 to aletterbox or sidepanel format and then, delivered from the output unit780 as a corresponding format video output 425.

The second data stream D₂ fed from the second data stream output 759 tothe summer 778 is summed with the output of the divider 777 to a sumsignal which is then fed to the second input 531 of the second videodecoder 422. The sum signal is further transferred to a divider circuit531 while it is divided into three compressed forms of H_(L)V_(H),H_(H)V_(L), and H_(H)V_(H). The three compressed signals are then fed toa second 535, a third 536, and a fourth expander 537 respectively forconverting by expansion to H_(L)V_(H), H_(H)V_(L), and H_(H)V_(H) of theoriginal length. The three signals are summed with H_(L)V_(L) by thevideo mixer 548 to a composite HDTV signal which is fed through anoutput 546 of the second video decoder to the output unit 780. Finally,the HDTV signal is delivered from the output unit 780 as an HDTV videosignal 427.

The output unit 780 is arranged for detecting an error rate in thesecond data stream of the second data stream output 759 through an errorrate detector 782 and if the error rate is high, delivering H_(L)V_(L)of low resolution video data systematically.

Accordingly, the multi-level signal transmission system for digital TVsignal transmission and reception becomes feasible. For example, if a TVsignal transmitter station is near, both the first and second datastreams of a received signal can successfully be reproduced to exhibitan HDTV quality picture. If the transmitter station is far, the firstdata stream can be reproduced to H_(L)V_(L) which is converted to a lowresolution TV picture. Hence, any TV program will be intercepted in awider area and displayed at a picture quality ranging from HDTV to NTSClevel.

FIG. 66 is a block diagram showing another arrangement of the TVreceiver. As shown, the receiver unit 751 contains only a first datastream output 768 and thus, the processing of the second data stream orHDTV data is not needed so that the overall construction can beminimized. It is a good idea to have the first video decoder 421 shownin FIG. 31 as a video decoder of the receiver. Accordingly, an NTSClevel picture will be reproduced. The receiver is fabricated at muchless cost as having no capability to receive any HDTV level signal andwill widely be accepted in the market. In brief, the receiver can beused as an adapter tuner for interception of a digital TV signal withgiving no modification to the existing TV system including a display.

The TV receiver 781 may have a further arrangement shown in FIG. 67,which serves as both a satellite broadcast receiver for demodulation ofPSK signals and a terrestrial broadcast receiver for demodulation of ASKsignals. In action, a PSK signal received by a satellite antenna 32 ismixed by a mixer 786 with a signal from an oscillator 787 to a lowfrequency signal which is then fed through an input unit 34 to a mixer753 similar to one shown in FIG. 63. The low frequency signal of PSK orQAM mode in a given channel of the satellite TV system is transferred toa modulator 35 where two data streams D₁ and D₂ are reproduced from thesignal. D₁ and D₂ are sent through a divider 788 to a second videodecoder 422 where they are converted to a video signal which is thendelivered from an output unit 780. Also, a digital or analogueterrestrial TV signal intercepted by a terrestrial antenna 32 a is fedthrough an input unit 752 to the mixer 753 where one desired channel isselected by the same manner as described in FIG. 63 and detected to alow frequency base band signal. The signal of analogue form is sentdirectly to the demodulator 35 for demodulation. The signal of digitalform is then fed to a discrimination/reproducing circuit 757 where twodata streams D₁ and D₂ are reproduced from the signal. D₁ and D₂ areconverted by the second video decoder 422 to a video signal which isthen delivered further. A satellite analogue TV signal is transferred toa video demodulator 788 where it is AN modulated to an analogue videosignal which is then delivered from the output unit 780. As understood,the mixer 753 of the TV receiver 781 shown in FIG. 67 is arrangedcompatible between two, satellite and terrestrial, broadcast services.Also, a receiver-circuit including a detector 755 and an LPF 756 for AMmodulation of an analogue signal can be utilized compatible with adigital ASK signal of the terrestrial TV service. The major part of thearrangement shown in FIG. 67 is arranged for compatible use, thusminimizing a circuitry construction.

According to the embodiment, a 4-level ASK signal is divided into two,D₁ and D₂, level components for execution of the one-bit modemulti-level signal transmission. If an 8-level ASK signal is used asshown in FIG. 68, it can be transmitted in a one-bit mode three-level,D₁, D₂, and D₃, arrangement. A shown in FIG. 68, D₁ is assigned to eightsignal points 721 a, 721 b, 722 a, 722 b, 723 a, 723 b, 724 a, 724 b,each pair representing a two-bit pattern, D₂ is assigned to four smallsignal point groups 721, 722, 723, 724, each two groups representing atwo-bit pattern, and D₃ is assigned to two large signal point groups 725and 726 representing a two-bit pattern. More particularly, this isequivalent to a form in which each of the four signal points 721, 722,723, 724 shown in FIG. 57 is divided into two components thus producingthree different level data.

The three-level signal transmission is identical to that described inthe third embodiment and will no further be explained in detail.

In particular, the arrangement of the video encoder 401 of the thirdembodiment shown in FIG. 30 is replaced with a modification of whichblock diagram is FIG. 69. The operation of the modified arrangement issimilar and will no longer be explained in detail. Two video signaldivider circuits 404 and 404 a which may be sub-band filters areprovided forming a divider unit 794. The divider unit 794 may also bearranged more simple a shown in the block diagram of FIG. 70, in which asignal passes across one signal divider circuit two times at timedivision mode. More specifically, a video signal of e.g. HDTV or superHDTV from the input unit 403 is time-base compressed by a time-basecompressor 795 and fed to the divider circuit 404 where it is dividedinto four components, H_(H)V_(H)-H, H_(H)V_(L)-H, and H_(L)V_(H)-H, andH_(L)V_(L)-H at a first cycle. At the time, four switches 765, 765 a,765 b, 765 c remain turned to the position 1 so that H_(H)V_(H)-H,H_(H)V_(L)-H, and H_(L)V_(H)-H are transmitted to a compressing circuit405. Meanwhile, H_(L)V_(L)-H is fed back through the terminal 1 of theswitch 765 c to the time-base compressor 795. At a second cycle, thefour switches 765, 765 a, 765 b, 765 c turned to the position 2 and allthe four components of the divider circuit 404 are simultaneouslytransferred to the compressing circuit 405. Accordingly, the dividerunit 796 of FIG. 70 arranged for time division processing of an inputsignal can be constructed in a simpler dividing circuit form.

At the receiver side, such a video decoder as described in the thirdembodiment and shown in FIG. 30 is needed for three-level transmissionof a video signal. More particularly, a third video decoder 423 isprovided which contains two mixers 556 and 556 a of different processingcapability as shown in the block diagram of FIG. 71.

Also, the third video decoder 423 may be modified in which the sameaction is executed with one single mixer 556 as shown in FIG. 72. At thefirst timing, five switches 765, 765 a, 765 b, 765 c, 765 d remainsturned to the position 1. Hence, H_(L)V_(L), H_(L)V_(H), H_(H)V_(L), andH_(H)V_(H) are fed from a first 522, a second 522 a, a third 522 b and afourth expander 522 c to through their respective switches to the mixer556 where they are mixed to a single video signal. The video signalwhich represents H_(L)V_(L)-H of an input high resolution video signalis then fed back through the terminal 1 of the switch 765 d to theterminal 2 of the switch 765 c. At the second timing, the four switches765, 765 a, 765 b, 765 c are turned to the point 2. Thus, H_(H)V_(H)-H,H_(H)V_(L)-H, H_(L)V_(H)-H, and H_(L)V_(L)-H are transferred to themixer 556 where they are mixed to a single video signal which is thensent across the terminal 2 of the switch 765 d to the output unit 554for further delivery.

In this manner of time division processing of a three-level signal, twomixers can be replaced with one mixer.

More particularly, four components H_(L)V_(L), H_(L)V_(H), H_(H)V_(L),H_(H)V_(H) are fed to produce H_(L)V_(L)-H at the first timing. Then,H_(L)V_(L)-H, H_(H)V_(L)-H, and H_(H)V_(H)-H are fed at the secondtiming delayed from the first timing and mixed with H_(L)V_(L)-H to atarget video signal. It is thus essential to perform the two actions atan interval of time.

If the four components are overlapped each other or supplied in avariable sequence, they have to be time-base adjusted to a givensequence through using memories accompanied with their respectiveswitches 765, 765 a, 765 b, 765 c. In the foregoing manner, a signal istransmitted from the transmitter at two different timing periods asshown in FIG. 73 so that no time-base controlling circuit is needed inthe receiver which is thus arranged more compact.

As shown in FIG. 73, D₁ is the first data stream of a transmittingsignal and H_(L)V_(L), H_(L)V_(H), H_(H)V_(L), and H_(H)V_(H) aretransmitted on D₁ channel at the period of first timing. Then, at theperiod of second timing, H_(L)V_(H), H_(H)V_(L), and H_(H)V_(H) aretransmitted on D₂ channel. As the signal is transmitted in a timedivision sequence, the encoder in the receiver can be arranged moresimple.

The technique of reducing the number of the expanders in the decoderwill now be explained. FIG. 74(b) shows a time-base assignment of fourdata components 810, 810 a, 810 b, 810 c of a signal. When other fourdata components 811, 811 a, 811 b, 811 c are inserted between the fourdata components 811, 811 a, 811 b, 811 c respectively, the latter can betransmitted at intervals of time. In action, the second video decoder422 shown in FIG. 74(a) receives the four components of the first datastream D₁ at a first input 521 and transfers them through a switch 812to an expander 503 one after another. More particularly, the component810 first fed is expanded during the feeding of the component 811 andafter completion of processing the component 810, the succeedingcomponent 810 a is fed. Hence, the expander 503 can process a row of thecomponents at time intervals by the same time division manner as of themixer, thus substituting the simultaneous action of a number ofexpanders.

FIG. 75 is a time-base assignment of data components of an HDTV signal,in which H_(L)V_(L)(1) of an NTSC component of the first channel signalfor a TV program is allocated to a data domain 821 of D₁ signal. Also,H_(L)V_(H), H_(H)V_(L), and H_(H)V_(H) carrying HDTV additionalcomponents of the first channel signal are allocated to three domains821 a, 821 b, 821 c of D₂ signal respectively. There are provided otherdata components 822, 822 a, 822 b, 822 c between the data components ofthe first channel signal which can thus be expanded with an expandercircuit during transmission of the other data. Hence, all the datacomponents of one channel signal will be processed by a single expandercapable of operating at a higher speed.

Similar effects will be ensured by assignment of the data components toother domains 821, 821 a, 821 b, 821 c as shown in FIG. 76. This becomesmore effective in transmission and reception of a common 4 PSK or ASKsignal having no different digital levels.

FIG. 77 shows a time-base assignment of data components during physicaltwo-level transmission of three different signal level data: e.g. NTSC,HDTV, and super HDTV or low resolution NTSC, standard resolution NTSC,and HDTV. For example, for transmission of three data components of lowresolution NTSC, standard NTSC, and HDTV, the low resolution NTSC orH_(L)V_(L) is allocated to the data domain 821 of D₁ signal. Also,H_(L)V_(H), H_(H)V_(L), and H_(H)V_(H) of the standard NTSC componentare allocated to three domains 821 a, 821 b, 821 c respectively.H_(L)V_(H)-H, H_(H)V_(L)-H, and H_(H)V_(H)-H of the HDTV component areallocated to domains 823, 823 a, and 823 b respectively.

The foregoing assignment is associated with such a logic levelarrangement based on discrimination in the error correction capabilityas described in the second embodiment.

More particularly, H_(L)V_(L) is carried on D¹⁻¹ channel of the D₁signal. The D₁₋₁ channel is higher in the error correction capabilitythan D₁₋₂ channel, as described in the second embodiment. The D₁₋₁channel is higher in the redundancy but lower in the error rate than theD₁₋₂ channel and the date 821 can be reconstructed at a lower C/N ratethan that of the other data 821 a, 821 b, 821 c. More specifically, alow resolution NTSC component will be reproduced at a far location fromthe transmitter antenna or in a signal attenuating or shadow area, e.g.the interior of a vehicle. In view of the error rate, the data 821 ofD₁₋₁ channel is less affected by signal interference than the other data821 a, 821 b, 821 c of D₁₋₂ channel, while being specificallydiscriminated and stayed in a different logic level, as described in thesecond embodiment. While D₁ and D₂ are divided into two physicallydifferent levels, the levels determined by discrimination of thedistance between error correcting codes are arranged different in thelogic level.

The demodulation of D₂ data requires a higher C/N rate than that for D₁data. In action, H_(L)V_(L) or low resolution NTSC signal can at leastbe reproduced in a distant or lower C/N service area. H_(L)V_(H),H_(H)V_(L), and H_(H)V_(H) can in addition be reproduced at a lower C/Narea. Then, at a high C/N area, H_(L)V_(H)-H, H_(H)V_(L)-H, andH_(H)V_(H)-H components can also be reproduced to develop an HDTVsignal. Accordingly, three different level broadcast signals can beplayed back. This method allows the signal receivable area shown in FIG.53 to increase from a double region to a triple region, as shown in FIG.90, thus ensuring higher opportunity for enjoying TV programs

FIG. 78 is a block diagram of the third video decoder arranged for thetime-base assignment of data shown in FIG. 77, which is similar to thatshown in FIG. 72 except that the third input 551 for D₃ signal iseliminated and the arrangement shown in FIG. 74(a) is added.

In operation, both the D₁ and D₂ signals are fed through two input units521, 530 respectively to a switch 812 at the first timing. As theircomponents including H_(L)V_(L) are time divided, they are transferredin a sequence by the switch 812 to an expander 503. This sequence willnow be explained referring to the time-base assignment of FIG. 77. Acompressed form of H_(L)V_(L) of the first channel is first fed to theexpander 503 where it is expanded. Then, H_(L)V_(H), H_(H)V_(L), andH_(H)V_(H) are expanded. All the four expanded components are sentthrough a switch 812 a to a mixer 556 where they are mixed to produceH_(L)V_(L)-H. H_(L)V_(L)-H is then fed back from the terminal 1 of aswitch 765 a through the input 2 of a switch 765 to the H_(L)V_(L) inputof the mixer 556.

At the second timing, H_(L)V_(H)-H, H_(H)V_(L)-H, and H_(H)V_(H)-H ofthe D₂ signal shown in FIG. 77 are fed to the expander 503 where theyare expanded before transferred through the switch 821 a to the mixer556. They are mixed by the mixer 556 to an HDTV signal which is fedthrough the terminal 2 of the switch 765 a to the output unit 521 forfurther delivery. The time-base assignment of data components fortransmission, shown in FIG. 77, contributes to the simplest arrangementof the expander and mixer. Although FIG. 77 shows two, D₁ and D₂, signallevels, four-level transmission of a TV signal will be feasible usingthe addition of a D₃ signal and a super resolution HDTV signal.

FIG. 79 illustrates a time-base assignment of data components of aphysical three-level, D₁, D₂, D₃, TV signal, in which data components ofthe same channel are so arranged as not to overlap with one another withtime. FIG. 80 is a block diagram of a modified video decoder 423,similar to FIG. 78, in which a third input 521 a is added. The time-baseassignment of data components shown in FIG. 79 also contributes to thesimple construction of the decoder.

The action of the modified decoder 423 is almost identical to that shownin FIG. 78 and associated with the time-base assignment shown in FIG. 77and will no more be explained. It is also possible to multiplex datacomponents on the D₁ signal as shown in FIG. 81. However, two data 821and 822 are increased higher in the error correction capability thanother data components 821 a, 812 b, 812 c, thus staying at a highersignal level. More particularly, the data assignment for transmission ismade in one physical level but two logic level relationship. Also, eachdata component of the second channel is inserted between two adjacentdata components of the first channel so that serial processing can beexecuted at the receiver side and the same effects as of the time-baseassignment shown in FIG. 79 will thus be obtained.

The time-base assignment of data components shown in FIG. 81 is based onthe logic level mode and can also be carried in the physical level modewhen the bit transmission rate of the two data components 821 and 822 isdecreased to ½ or ⅓ thus to lower the error rate. The physical levelarrangement is consisted of three different levels.

FIG. 82 is a block diagram of another modified video decoder 423 fordecoding of the D₁ signal time-base arranged as shown in FIG. 81, whichis simpler in construction than that shown in FIG. 80. Its action isidentical to that of the decoder shown in FIG. 80 and will be no moreexplained.

As understood, the time-base assignment of data components shown in FIG.81 also contributes to the similar arrangement of the expander andmixer. Also, four data components of the D₁ signal are fed at respectivetime slices to a mixer 556. Hence, the circuitry arrangement of themixer 556 or a plurality of circuit blocks such as provided in the videomixer 548 of FIG. 32 may be arranged for changing the connectiontherebetween corresponding to each data component so that they becomecompatible in time division action and thus, minimized in circuitryconstruction.

Accordingly, the receiver can be minimized in the overall construction.

It would be understood that the fifth embodiment is not limited to ASKmodulation and the other methods including PSK and QAM modulation, suchas described in the first, second, and third embodiments, will beemployed with equal success.

Also, FSK modulation will be eligible in any of the embodiments. Forexample, the signal points of a multiple-level FSK signal consisting offour frequency components f1, f2, f3, f4 are divided into groups asshown in FIG. 58 and when the distance between any two groups are spacedfrom each other for ease of discrimination, the multi-level transmissionof the FSK signal can be implemented, as illustrated in FIG. 83.

More particularly, it is assumed that the frequency group 841 of f1 andf2 is assigned D₁=0 and the group 842 of f3 and f4 is assigned D₁=1. Iff1 and f3 represent 0 at D₂ and f2 and f4 represent 1 at D₂, two-bitdata transmission, one bit at D₁ or D₂, will be possible as shown inFIG. 83. When the C/N rate is high, a combination of D₁=0 and D₁=1 isreconstructed at t=t3 and a combination of D₁=1 and D₂=0 at t=t4. Whenthe C/N rate is low, D₁=0 only is reproduced at t=t3 and D₁=1 at t=t4.In this manner, the FSK signal can be transmitted in the multi-levelarrangement. This multi-state FSK signal transmission is applicable toeach of the third, fourth, and fifth embodiments.

The fifth embodiment may also be implemented in the form of a magneticrecord/playback apparatus of which block diagram shown in FIG. 84because its ASK mode action is appropriate to magnetic record andplayback operation.

Embodiment 6

A sixth embodiment of the present invention is applicable to a magneticrecording and playback apparatus. Although the present invention isapplied for a multiple-level recording ASK data transmission in theabove-described fifth embodiment, it is also feasible in the same mannerto adopt this invention in a magnetic recording and playback apparatusof a multi-level ASK recording system. A multi-level magnetic recordingcan be realized by applying the C-CDM method of the present invention toPSK, FCK, and QAM, as well as ASK.

First of all, the method of realizing a multi-level recording in a 16QAM or 32 QAM magnetic recording playback apparatus will be explained incompliance with the C-CDM method of the present invention. FIG. 84 is acircuit block diagram showing a QAM system incorporating C-CDMmodulator. Hereinafter, a QAM system being multiplexed by the C-CDMmethod is termed as SRQAM.

As shown in FIG. 84, an input video signal, e.g. an HDTV signal, to amagnetic record/playback apparatus 851 is divided and compressed by avideo encoder 401 into a low frequency band signal through a first videoencoder 401 a and a high frequency band signal through a second videoencoder 401 b respectively. Then, a low frequency band component, e.g.H_(L)V_(L), of the video signal is fed to a first data stream input 743of an input unit 742 and a high frequency band component includingH_(H)V_(H) is fed to a second data stream input 744 of the same. The twocomponents are further transferred to a modulator 749 of amodulator/demodulator unit 852. The first data stream input 743 adds anerror correcting code to the low frequency band signal in an ECC 743 a.On the other hand, the second data stream fed into the second datastream input 744 is 2 bit in case of 16 SRQAM, 3 bit in case of 36SRQAM, and 4 bit in case of 64 SRQAM. After an error correcting codebeing encoded by an ECC 744 a, this signal is supplied to a Trellisencoder 744 b in which a Trellis encoded signal having a ratio 1/2 incase of 16 SRQAM, 2/3 in case of 32 SRQAM, and 3/4 in case of 64 SRQAMis produced. A 64 SRQAM signal, for example, has a first data stream of2 bit and a second data stream of 4 bit. A Trellis encoder of FIG. 128allows this 64 SRQAM signal to perform a Trellis encoding of ratio 3/4wherein 3 bit data is converted into 4 bit data. Thus redundancyincreases and a data rate decreases, while error correcting capabilityincreases. This results in the reduction of an error rate in the samedata rate. Accordingly, transmittable information amount of therecording/playback system or transmission system will increasesubstantially.

It is, however, possible to constitute the first data stream input 743not to include a Trellis encoder as shown in FIG. 84 of this sixthembodiment because the first data stream has low error rate inherently.This will be advantageous in view of simplification of circuitconfiguration. The second data stream, however, has a narrow inter-codedistance as compared with the first data stream and, therefore, has aworse error rate. The Trellis encoding of the second data streamimproves such a worse error rate. It is no doubt that an overall circuitconfiguration becomes simple if the Trellis encoding of the first datastream is eliminated. An operation for modulation is almost identical tothat of the transmitter of the fifth embodiment shown in FIG. 64 andwill be no more explained. A modulated signal of the modulator 749 isfed into a recording/playback circuit 853 in which it is AC biased by abias generator 856 and amplified by an amplifier 857 a. Thereafter, thesignal is fed to a magnetic head 854 for recording onto a magnetic tape855.

A format of the recorded signal is shown in a recording signal frequencyassignment of FIG. 113. A main, e.g. 16 SRQAM, signal 859 having acarrier of frequency fc records information, and also a pilot f_(p)signal 859 a having a frequency 2 fc is recorded simultaneously.Distortion in the recording operation is lowered as a bias signal 859 bhaving a frequency f_(BIAS) adds AC bias for magnetic recording. Two ofthree-level signals shown in FIG. 113 are recorded in multiple state. Inorder to reproduce these recorded signals, two thresholds Th-1-2, Th-2are given. A signal 859 will reproduce all of two levels while a signal859 c will reproduce D₁ data only, depending on C/N level of therecording/playback.

A main signal of 16 SRQAM will have a signal point assignment shown inFIG. 10. Furthermore, a main signal of 36 SRQAM will have a signal pointassignment shown in FIG. 100. In reproduction of this signal, both themain signal 859 and the pilot signal 859 a are reproduced through themagnetic head 854 and amplified by an amplifier 857 b. An output signalof the amplifier 857 b is fed to a carrier reproduction circuit 858 inwhich a filter 858 a separates the frequency of the pilot signal f_(p)having a frequency 2 f 0 and a ½ frequency divider 858 b reproduces acarrier of frequency f0 to transfer it to a demodulator 760. Thisreproduced carrier is used to demodulate the main signal in thedemodulator 760. Assuming that a magnetic recording tape 855, e.g. HDTVtape, is of high C/N rate, 16 signal points are discriminatable and thusboth D₁ and D₂ are demodulated in the demodulator 760. Subsequently, avideo decoder 402 reproduce all the signals. An HDTV VCR can reproduce ahigh bit-rate TV signal such as 15 Mbps HDTV signal. The low the C/Nrate is, the cheaper the cost of a video tape is. So far, a VHS tape inthe market is inferior more than 10 dB in C/N rate to a full-scalebroadcast tape. If a video tape 855 is of low C/N rate, it will not beable to discriminate all the 16 or 32 valued signal points. Thereforethe first data stream D₁ can be reproduced, while a 2 bit, 3 bit, or 4bit data stream of the second data stream D₂ cannot be reproduced. Only2 bit data stream of the first data stream is reproduced. If a two-levelHDTV video signal is recorded and reproduced, a low C/N tape havinginsufficient capability of reproducing a high frequency band videosignal can output only a low rate low frequency band video signal of thefirst data stream, specifically e.g. a 7 Mbps wide NTSC TV signal.

As shown in a block diagram of FIG. 114, a second data stream output759, the second data stream input 744, and the second video decoder 402a can be eliminated in order to provide customers one aspect of lowergrade products. In this case, a recording/playback apparatus 851,dedicated to a low bit rate, will include a modulator such as amodulated QPSK which modulates or demodulates the first data streamonly. This apparatus allows only the first data stream to be recordedand reproduced. Specifically, a wide NTSC grade video signal can berecorded and reproduced.

Above-described high C/N rate video tape 855 capable of recording a highbit-rate signal, e.g. HDTV signal, will be able to use in such a lowbit-rate dedicated magnetic recording/playback apparatus but willreproduce the first data stream D₁ only. That is, the wide NTSC signalis outputted, while the second data stream is not reproduced. In otherwords, one recording/playback apparatus having a complicatedconfiguration can reproduce a HDTV signal and the otherrecording/playback apparatus having a simple configuration can reproducea wide NTSC signal if a given video tape 855 includes the samemulti-level HDTV signal. Accordingly in case of two-level multiplestate, four combinations will be realized with perfect compatibilityamong two tapes having different C/N rates and two recording/playbackapparatus having different recording/playback data rates. This willbring remarkable effect. In this case, an NTSC dedicated apparatus willbe simple in construction as compared with an HDTV dedicated apparatus.In more detail, a circuitry scale of EDTV decoder will be ⅙ of that ofHDTV decoder. Therefore, a low function apparatus can be realized atfairly low cost. Realization of two, HDTV and EDTV, typesrecording/playback apparatus having different recording/reproducingcapability of picture quality will provide various type products rangingin a wide price range. Users can freely select a tape among a pluralityof tapes from an expensive high C/N rate tape to a cheaper low C/N ratetape, as occasion demands so as to satisfy required picture quality. Notonly maintaining perfect compatibility but obtaining expandablecapability will be attained and further compatibility with a futuresystem will be ensured. Consequently, it will be possible to establishlong-lasting standards for recording/playback apparatus. Other recordingmethods will be used in the same manner. For example, a multi-levelrecording will be realized by use of phase modulation explained in thefirst and third embodiments. A recording using ASK explained in thefifth embodiment will also be possible. A multiple state will berealized by converting present recording from two-level to four-leveland dividing into two groups as shown in FIGS. 59(c) and 59(d).

A circuit block diagram for ASK is identical to that disclosed in FIG.84. Besides embodiments already described, a multi-level recording willbe also realized by use of multiple tracks on a magnetic tape.Furthermore, a theoretical multi-level recording will be feasible bydifferentiating the error correcting capability so as to discriminaterespective data.

Compatibility with future standards will be described below. A settingof standards for recording/playback apparatus such as VCR is normallydone by taking account of the most highest C/N rate tape available inpractice. The recording characteristics of a tape progresses rapidly.For example, the C/N rate has been improved more than 10 dB comparedwith the tape used 10 years ago. If supposed that new standards will beestablished after 10 to 20 years due to an advancement of tape property,a conventional method will encounter with difficulty in maintainingcompatibility with older standards. New and old standards, in fact, usedto be one-way compatible or non-compatible with each other. On thecontrary, in accordance with the present invention, the standards arefirst of all established for recording and/or reproducing the first datastream and/or second data stream on present day tapes. Subsequently, ifthe C/N rate is improved magnificently in future, an upper level datastream, e.g. a third data stream, will be added without any difficultyas long as the present invention is incorporated in the system. Forexample, a super HDTV VCR capable of recording or reproducingthree-level 64 SRQAM will be realized while maintaining perfectcompatibility with the conventional standards. A magnetic tape,recording first to third data streams in compliance with new standards,will be able to use, of course, in the older two-level magneticrecording/playback apparatus capable of recording and/or reproducingonly first and second data streams. In this case, first and second datastreams can be reproduced perfectly although the third data stream isleft non-reproduced. Therefore, an HDTV signal can be reproduced. Forthese reasons, the merit of expanding recording data amount whilemaintaining compatibility between new and old standards is expected.

Returning to the explanation of reproducing operation of FIG. 84, themagnetic head 854 and the magnetic reproduction circuit 853 reproduce areproducing signal from the magnetic tape 855 and feeds it to themodulation/demodulation circuit 852. The demodulating operation isalmost identical with that of first, third, and fourth embodiments andwill no further be explained. The demodulator 760 reproduces the firstand second data streams D₁ and D₂. The second data stream D₂ is errorcorrected with high code gain in a Trellis-decoder 759 b such as aVitabi decoder, so as to be low error rate. The video decoder 402demodulates D₁ and D₂ signals to output an HDTV video signal.

FIG. 131 is a block diagram showing a three-level magneticrecording/playback apparatus in accordance with the present inventionwhich includes one theoretical level in addition to two physical levels.This system is substantially the same as that of FIG. 84. The differenceis that the first data stream is further divided into two subchannels byuse of a TDM in order to realize a three-level constitution.

As shown in FIG. 131, an HDTV signal is separated first of all into two,medium and low frequency band video signals D₁₋₁ and D₁₋₂, through a 1-1video encoder 401 c and a 1-2 video encoder 401 d and, thereafter, fedinto a first data stream input 743 of an input section 742. The datastream D₁₋₁ having a picture quality of MPEG grade is error correctingcoded with high code gain in an ECC coder 743 a, while the data streamD₁₋₂ is error correcting coded with normal code gain in an ECC encoder743 b. D₁₋₁ and D₁₋₂ are time multiplexed together in a TDM 743 c to beone data stream D1. D₁ and D₂ are modulated into two-level signals in aC-CDM 749 and then recorded on the magnetic tape 855 through themagnetic head 854.

In playback operation, a recording signal reproduced through themagnetic head 854 is demodulated into D₁ and D₂ by the C-CDM demodulator760 in the same manner as in the explanation of FIG. 84. The first datastream D₁ is demodulated into two, D₁₋₁ and D₁₋₂, subchannels throughthe TDM 758 c provided in the first data stream output 758. D₁₋₁ data iserror corrected in an ECC decoder 758 a having high code gain.Therefore, D₁₋₁ data can be demodulated at a lower C/N rate as comparedwith D₁₋₂ data. A 1-1 video decoder 402 a decodes the D₁₋₁ data andoutputs an LDTV signal. On the other hand, D₁₋₂ data is error correctedin an ECC decoder 758 b having normal code gain. Therefore, D₁₋₂ datahas a threshold value of high C/N rate compared with D₁₋₁ data and thuswill not be demodulated when a signal level is not large. D₁₋₂ data isthen demodulated in a 1-2 video decoder 402 d and summed with D₁₋₁ datato output an EDTV signal of wide NTSC grade.

The second data stream D₂ is Vitabi demodulated in a Trellis decoder 759b and error corrected at an ECC decoder 759 a. Thereafter, D₂ data isconverted into a high frequency band video signal through a second videodecoder 402 b and, then, summed with D₁₋₁ and D₁₋₂ data to output anHDTV signal. In this case, a threshold value of the C/N rate of D₂ datais set larger than that of C/N rate of D₁₋₂ data. Accordingly, D₁₋₁data, i.e. an LDTV signal, will be reproduced from a tape 855 having asmaller C/N rate. D₁₋₁ and D₁₋₂ data, i.e. an EDTV signal, will bereproduced from a tape 855 having a normal C/N rate. And, D₁₋₁, D₁₋₂,and D₂, i.e. an HDTV signal, will be reproduced from a tape 855 having ahigh C/N rate.

Three-level magnetic recording/playback apparatus can be realized inthis manner. As described in the foregoing description, the tape 855 hasan interrelation between C/N rate and cost. The present invention allowsusers to select a grade of tape in accordance with a content of TVprogram they want to record because video signals having picturequalities of three grades are recorded and/or reproduced in accordancewith tape cost.

Next, an effect of multi-level recording will be described with respectto fast feed playback. As shown in a recording track diagram of FIG.132, a recording track 855 a having an azimuth angle A and a recordingtrack 855 b having an opposite azimuth angle B are alternately arrayedon the magnetic tape 855. The recording track 855 a has a recordingregion 855 c at its central portion and the remainder as D₁₋₂ recordingregions 855 d, as denoted in the drawing. This unique recording patternis provided on at least one of several recording tracks. The recordingregion 855 c records one frame of LDTV signal. A high frequency bandsignal D₂ is recorded on a D₂ recording region 855 e corresponding to anentire recording region of the recording track 855 a. This recordingformat causes no novel effect against a normal speed recording/playbackoperation.

A fast feed reproduction in a reverse direction does not allow amagnetic head trace 855 f having an azimuth angle A to coincide with themagnetic track as shown in the drawing. As the present inventionprovides the D₁₋₁ recording region 855 c at a central narrow region ofthe magnetic tape as shown in FIG. 132, this region only is surelyreproduced although it occurs at a predetermined probability. Thusreproduced D₁₋₁ signal can demodulate an entire picture plane of thesame time although its picture quality is an LDTV of MPEG1 level. Inthis manner several to several tens LDTV signals per second can bereproduced with perfect picture images during the fast feed playbackoperation, thereby enabling users to surely confirm picture imagesduring the fast feed operation.

A head trace 855 g corresponds to a head trace in the reverse playbackoperation, from which it is understood only a part of the magnetic trackis traced in the reverse playback operation. The recording/playbackformat shown in FIG. 132 however allows, even in such a reverse playbackoperation, to reproduce D₁₋₁ recording region and, therefore, ananimation of LDTV grade is outputted intermittently.

Accordingly, the present invention makes it possible to record a pictureimage of LDTV grade within a narrow region on the recording track, whichresults in intermittent reproduction of almost perfect still pictureswith picture quality of LDTV grade during normal and reverse fast feedplayback operations. Thus, the users can easily confirm picture imageseven in high-speed searching.

Next, another method will be described to respond a higher speed fastfeed playback operation. A D₁₋₁ recording region 855 c is provided asshown at lower right of FIG. 132, so that one frame of LDTV signal isrecorded thereon. Furthermore, a narrow D₁₋₁·D₂ recording region 855 his provided at a part of the D₁₋₁ recording region 855 c. A subchannelD₁₋₁ in this region records a part of information relating to the oneframe of LDTV signal. The remainder of the LDTV information is recordedon the D₂ recording region 855 j of the D₁₋₁·D₂ recording region 855 hin a duplicated manner. The subchannel D₁₋₁ has a data recordingcapacity 3 to 5 times as much as the subchannel D₁₋₁. Therefore,subchannels D₁₋₁ and D₂ can record one frame information of LDTV signalon a smaller, ⅓˜⅕, area of the recording tape. As the head trace can berecorded in a further narrower regions 855 h, 855 j, both time and areaare decreased into ⅓˜⅕ as compared with a head trace time T_(S1). Evenif the trace of head is further inclined by increasing fast feed speedamount, the probability of entirely tracing this region will beincreased. Accordingly, perfect LDTV picture images will beintermittently reproduced even if the fast feed speed is increased up to3 to 5 times as fast as the case of the subchannel D₁₋₁ only.

In case of a two-level VCR, this method is useless in reproducing the D₂recording region 855 j and therefore this region will not be reproducedin a high-speed fast feed playback operation. On the other hand, athree-level high performance VCR will allow users to confirm a pictureimage even if a fast feed playback operation is executed at a faster, 3to 5 times as fast as two-level VCR, speed. In other words, not onlyexcellent picture quality is obtained in accordance with the cost but amaximum fast feed speed capable of reproducing picture images can beincreased in accordance with the cost.

Although this embodiment utilizes a multi-level modulation system, it isneedless to say that a normal, e.g. 16 QAM, modulation system can alsobe adopted to realize the fast feed playback operation in accordancewith the present invention as long as an encoding of picture images isof multiple type.

A recording method of a conventional non-multiple digital VCR, in whichpicture images are highly compressed, disperses video data uniformly.Therefore, it was not possible in a fast feed playback operation toreproduce all the picture images on a picture plane of the same time.The picture reproduced was the one consisting of a plurality of pictureimage blocks having non-coincided time bases with each other. Thepresent invention, however, provides a multi-level HDTV VCR which canreproduce picture image blocks having coincided time bases on a pictureplane during a fast feed playback operation although its picture qualityis of LDTV grade.

The three-level recording in accordance with the present invention willbe able to reproduce a high resolution TV signal such as HDTV signalwhen the recording/playback system has a high C/N rate. Meanwhile, a TVsignal of EDTV grade, e.g. a wide NTSC signal, or a TV signal of LDTVgrade, e.g. a low resolution NTSC signal, will be outputted when therecording/playback system has a low C/N rate or poor function.

As is described in the foregoing description, the magneticrecording/playback apparatus in accordance with the present inventioncan reproduce picture images consisting of the same content even if C/Nrate is low or error rate is high, although the resolution or thepicture quality is relatively low.

Embodiment 7

A seventh embodiment of the present invention will be described forexecution of four-level video signal transmission. A combination of thefour-level signal transmission and the four-level video dataconstruction will create a four-level signal service area as shown inFIG. 91. The four-level service area is consisted of, from innermost, afirst 890 a, a second 890 b, a third 890 c, and a fourth signalreceiving area 890 d. The method of developing such a four-level servicearea will be explained in more detail.

The four-level arrangement can be implemented by using four physicallydifferent levels, determined through modulation or four logic levelsdefined by data discrimination in the error correction capability. Theformer provides a large difference in the C/N rate between two adjacentlevels and the C/N rate has to be increased to discriminate all the fourlevels from each other. The latter is based on the action ofdemodulation and a difference in the C/N rate between two adjacentlevels should stay at minimum. Hence, the four-level arrangement is bestconstructed using a combination of two physical levels and two logiclevels. The division of a video signal into four signal levels will beexplained.

FIG. 93 is a block diagram of a divider circuit 3 which comprises avideo divider 895 and four compressors 405 a, 405 b, 405 c, 405 d. Thevideo divider 895 contains three dividers 404 a, 404 b, 404 c which arearranged identical to the divider circuit 404 of the first video encoder401 shown in FIG. 30 and will be no more explained. An input videosignal is divided by the dividers into four components, H_(L)V_(L) oflow resolution data, H_(H)V_(H) of high resolution data, and H_(L)V_(H)and H_(H)V_(L) for medium resolution data. The resolution of H_(L)V_(L)is a half that of the original input signal.

The input video signal is first divided by the divider 404 a into two,high and low, frequency band components, each component being dividedinto two, horizontal and vertical, segments. The intermediate betweenthe high and low frequency ranges is a dividing point according to theembodiment. Hence, if the input video signal is an HDTV signal of1000-line vertical resolution, H_(L)V_(L) has a vertical resolution of500 lines and a horizontal resolution of a half value.

Each of two, horizontal and vertical, data of the low frequencycomponent H_(L)V_(L) is further divided by the divider 404 c into twofrequency band segments. Hence, an H_(L)V_(L) segment output is 250lines in the vertical resolution and ¼ of the original horizontalresolution. This output of the divider 404 c which is termed as an LLsignal is then compressed by the compressor 405 a to a D₁₋₁ signal.

The other three higher frequency segments of H_(L)V_(L) are mixed by amixer 772 c to an LH signal which is then compressed by the compressor405 b, to a D₁₋₂ signal. The compressor 405 b may be replaced with threecompressors provided between the divider 404 c and the mixer 772 c.

H_(L)V_(H), H_(H)V_(L), and H_(H)V_(H) form the divider 404 a are mixedby a mixer 772 a to an H_(H)V_(H)-H signal. If the input signal is ashigh as 1000 lines in both horizontal and vertical resolution,H_(H)V_(H)-H has 500 to 1000 lines of a horizontal and a verticalresolution. H_(H)V_(H)-H is fed to the divider 404 b where it is dividedagain into four components.

Similarly, H_(L)V_(L) from the divider 404 b has 500 to 750 lines of ahorizontal and a vertical resolution and transferred as an HL signal tothe compressor 405 c. The other three components, H_(L)V_(H),H_(H)V_(L), and H_(H)V_(H), from the divider 404 b have 750 to 1000lines of a horizontal and a vertical resolution and are mixed by a mixer772 b to an HH signal which is then compressed by the compressor 405 dand delivered as a D₂₀₂ signal. After compression, the HL signal isdelivered as a D₂₋₁ signal. As the result, LL or D₁₋₁ carries afrequency data of 0 to 250 lines, LH or D₁₋₂ carries a frequency datafrom more than 250 lines up to 500 lines, HL or D₂₋₁ carries a frequencydata of more than 500 lines up to 750 lines, and HH or D₂₋₂ carries afrequency data of more than 750 lines to 1000 lines so that the dividercircuit 3 can provide a four-level signal. Accordingly, when the dividercircuit 3 of the transmitter 1 shown in FIG. 87 is replaced with thedivider circuit of FIG. 93, the transmission of a four-level signal willbe implemented.

The combination of multi-level data and multi-level transmission allowsa video signal to be at steps declined in the picture quality inproportion to the C/N rate during transmission, thus contributing to theenlargement of the TV broadcast service area. At the receiving side, theaction of demodulation and reconstruction is identical to that of thesecond receiver of the second embodiment shown in FIG. 88 and will be nomore explained. In particular, the mixer 37 is modified for video signaltransmission rather than data communications and will now be explainedin more detail.

As described in the second embodiment, a received signal afterdemodulated and error corrected, is fed as a set of four componentsD₁₋₁, D₁₋₂, D₂₋₁, D₂₋₂ to the mixer 37 of the second receiver 33 of FIG.88.

FIG. 94 is a block diagram of a modified mixer 33 in which D₁₋₁, D₁₋₂,D₂₋₁, D₂₋₂ are explained by their respective expanders 523 a, 523 b, 523c, 523 d to an LL, and LH, an HL, and an HH signal respectively whichare equivalent to those described with FIG. 93. If the bandwidth of theinput signal is 1, LL has a bandwidth of ¼, LL+LH has a bandwidth of ½,LL+LH+HL has a bandwidth of ¾, and LL+LH+HL+HH has a bandwidth of 1. TheLH signal is then divided by a divider 531 a and mixed by a video mixer548 a with the LL signal. An output of the video mixer 548 a istransferred to an H_(L)V_(L) terminal of a video mixer 548 c. The videomixer 531 a is identical to that of the second decoder 527 of FIG. 32and will be no more explained. Also, the HH signal is divided by adivider 531 b and fed to a video mixer 548 b. At the video mixer 548 b,the HH signal is mixed with the HL signal to an H_(H)V_(H)-H signalwhich is then divided by a divider 531 c and sent to the video mixer 548c. At the video mixer 548 c, H_(H)V_(H)-H is combined with the sumsignal of LH and LL to a video output. The video output of the mixer 33is then transferred to the output unit 36 of the second receiver shownin FIG. 88 where it is converted to a TV signal for delivery. If theoriginal signal has 1050 lines of vertical resolution or is an HDTVsignal of about 1000-line resolution, its four different signal levelcomponents can be intercepted in their respective signal receiving areasshown in FIG. 91.

The picture quality of the four different components will be describedin more detail. The illustration of FIG. 92 represents a combination ofFIGS. 86 and 91. As apparent, when the C/N rate increases, the overallsignal level of amount of data is increased from 862 d to 862 a by stepsof four signal levels D₁₋₁, D₁₋₂, D₂₋₁, D₂₋₂.

Also, as shown in FIG. 95, the four different level components LL, LH,HL, and HH are accumulated in proportion to the C/N rate. Morespecifically, the quality of a reproduced picture will be increased asthe distance from a transmitter antenna becomes small. When L=Ld, LLcomponent is reproduced. When L=Lc, LL+LH signal is reproduced. WhenL=Lb, LL+LH+HL signal is reproduced. When L=La, LL+LH+HL+HH signal isreproduced. As the result, if the bandwidth of the original signal is 1,the picture quality is enhanced at ¼ increments of bandwidth from ¼ to 1depending on the receiving area. If the original signal is an HDTV of1000-line vertical resolution, a reproduced TV signal is 250, 500, 750,and 1000 lines in the resolution at their respective receiving areas.The picture quality will thus be varied at steps depending on the levelof a signal. FIG. 96 shows the signal propagation of a conventionaldigital HDTV signal transmission system, in which no signal reproductionwill be possible when the C/N rate is less than V0. Also, signalinterception will hardly be guaranteed at signal interference regions,shadow regions, and other signal attenuating regions, denoted by thesymbol x, of the service area. FIG. 97 shows the signal propagation ofan HDTV signal transmission system of the present invention. As shown,the picture quality will be a full 1000-line grade at the distance Lawhere C/N=a, a 750-line grade at the distance Lb where C/N=b, a 500-linegrade at the distance Lc where C/N=c, and a 250-line grade at thedistance Ld where C/N=d. Within the distance La, there are shownunfavorable regions where the C/N rate drops sharply and no HDTV qualitypicture will be reproduced. As understood, a lower picture qualitysignal can however be intercepted and reproduced according to themulti-level signal transmission system of the present invention. Forexample, the picture quality will be a 750-line grade at the point B ina building shadow area, a 250-line grade at the point D in a runningtrain, a 750-line grade at the point F in a ghost developing area, a250-line grade at the point G in a running car, a 250-line grade at thepoint L in a neighbor signal interference area. As set forth above, thesignal transmission system of the present invention allows a TV signalto be successfully received at a grade in the area where theconventional system is poorly qualified, thus increasing its servicearea. FIG. 98 shows an example of simultaneous broadcasting of fourdifferent TV programs, in which three quality programs C, B, A aretransmitted on their respective channels D₁₋₂, D₂₋₁, D₂₋₂ while aprogram D identical to that of a local analogue TV station is propagatedon the D₁₋₁ channel. Accordingly, while the program D is kept availableat simulcast service, the other three programs can also be distributedon air for offering a multiple program broadcast service.

Embodiment 8

Hereinafter, an eighth embodiment of the present invention will beexplained referring to the drawings. The eighth embodiment employs amulti-level signal transmission system of the present invention for atransmitter/receiver in a cellular telephone system.

FIG. 115 is a block diagram showing a transmitter/receiver of a portabletelephone, in which a telephone conversation sound inputted across amicrophone 762 is compressed and coded in a compressor 405 intomulti-level, D₁, D₂, and D₃, data previously described. These D₁, D₂,and D₃ data are time divided in a time division circuit 765 intopredetermined time slots and, then, modulated in a modulator 4 into amulti-level, e.g. SRQAM, signal previously described. Thereafter, anantenna sharing unit 764 and an antenna 22 transmit a carrier wavecarrying a modulated signal, which will be intercepted by a base stationlater described and further transmitted to other base stations or acentral telephone exchanger so as to communicate with other telephones.

On the contrary, the antenna 22 receives transmission radio waves fromother base stations as communication signals from other telephones. Areceived signal is demodulated in a multiple-level, e.g. SRQAM, typedemodulator 45 into D₁, D₂, and D₃ data. A timing circuit 767 detectstiming signals on the basis of demodulated signals. These timing signalsare fed into the time division circuit 765. Demodulated signals D₁, D₂,and D₃ are fed into an expander 503 and expanded into a sound signal,which are transmitted to a speaker 763 and converted into sound.

FIG. 116 shows a block diagram exemplarily showing an arrangement ofbase stations, in which three base stations 771, 772, and 773 locate atcenter of respective receiving cells 768, 769, and 770 of hexagon orcircle. These base stations 771, 772, and 773 respectively has aplurality of transmitter/receiver units 761 a˜761 j each similar to thatof FIG. 115 so as to have data communication channels equivalent to thenumber of these transmitter/receiver units. A base station controller774 is connected to all the base stations and always monitors acommunication traffic amount of each base station. Based on themonitoring result, the base station controller 774 carries out anoverall system control including allocation of channel frequencies torespective base stations or control of receiving cells of respectivebase stations.

FIG. 117 is a view showing a traffic distribution of communicationamount in a conventional, e.g. QPSK, system. A diagram d=A shows data774 a and 774 b having frequency utilization efficiency 2 bit/Hz, and adiagram d=B shows data 774 c of frequency utilization efficiency 2bit/Hz. A summation of these data 774 a, 774 b, and 774 c becomes a data774 d, which represents a transmission amount of Ach consisting ofreceiving cells 768 and 770. Frequency utilization efficiency of 2bit/Hz is uniformly distributed. However, density of population in anactual urban area is locally high in several crowded areas 775 a, 775 b,and 775 c which includes buildings concentrated. A data 774 erepresenting a communication traffic amount shows several peaks atlocations just corresponding to these crowded areas 775 a, 775 b, and775 c, in contrast with other area having small communication amount. Acapacity of a conventional cellular telephone was uniformly set to 2bit/Hz frequency efficiency at entire region as shown by the data 774 dirrespective of actual traffic amount TF shown by the data 774 e. It isnot effective to dive the same frequency efficiency regardless of actualtraffic amount. In order to compensate this ineffectiveness, theconventional systems have allocated many frequencies to the regionshaving a large traffic amount, increased channel number, or decreasedthe receiving cell of the same. However, an increase of channel numberis restricted by the frequency spectrum. Furthermore, conventionalmulti-level; e.g. 16 QAM or 64 QAM, mode transmission systems increasetransmission power. A reduction of receiving cell will induce anincrease in number of base stations, thus increasing installation cost.

It is ideal for the improvement of an overall system efficiency toincrease the frequency efficiency of the region having a larger trafficamount and decrease the frequency efficiency of the region having asmaller traffic amount. A multi-level signal transmission system inaccordance with the present invention realizes this ideal modification.This will be explained with reference to FIG. 118 showing acommunication amount & traffic distribution in accordance with theeighth embodiment of the present invention.

More specifically, FIG. 118 shows communication amounts of respectivereceiving cells 770 b, 768, 769, 770, and 770 a taken along a line A-A′.The receiving cells 768 and 770 utilize frequencies of a channel groupA, while the receiving cells 770 b, 769, and 770 a utilize frequenciesof a channel group B which does not overlap with the channel group A.The base station controller 774 shown in FIG. 116 increases or decreaseschannel number of these channels in accordance with the traffic amountof respective receiving cells. In FIG. 118, a diagram d=A represents adistribution of a communication amount of the A channel. A diagram d=Brepresents a distribution of a communication amount of the B channel. Adiagram d=A+B represents a distribution of a communication amount of allthe channels. A diagram TF represents a communication traffic amount,and a diagram P shows a distribution of buildings and population.

The receiving cells 768, 769, and 770 employ the multi-level, e.g.SRQAM, signal transmission system. Therefore, it is possible to obtain afrequency utilization efficiency of 6 bit/Hz, three times as large as 2bit/Hz of QPSK, in the vicinity of the base stations as denoted by data776 a, 776 b, and 776 c. Meanwhile, the frequency utilization efficiencydecreases at steps from 6 bit/Hz to 4 bit/Hz, and 4 bit/Hz to 2 bit/Hz,as it goes to suburban area. If the transmission power is insufficient,2 bit/Hz areas become narrower than the receiving cells, denoted bydotted lines 777 a, 777 b, 777 c, of QPSK. However, an equivalentreceiving cell will be easily obtained by slightly increasing thetransmission power of the base stations.

Transmitting/receiving operation of a mobile station capable ofresponding to a 64 SRQAM signal is carried out by use of modified QPSK,which is obtained by set a shift amount of SRQAM to S=1, at the placefar from the base station, by use of 16 SRQAM at a place not so far fromthe same, and 64 SRQAM at the near place. Accordingly, the maximumtransmission power does not increase as compared with QPSK.

Furthermore, 4 SRQAM type transmitter/receiver, whose circuitconfiguration is simplified as shown in a block diagram of FIG. 121,will be able to communicate with other telephones while maintainingcompatibility. That will be the same in 16 SRQAM typetransmitter/receiver shown in a block diagram of FIG. 122. As a result,three different type telephones having different modulation systems willbe provided. Small in size and light in weight is important for portabletelephones. In this regard, the 4 SRQAM system having a simple circuitconfiguration will be suitable for the users who want a small and lighttelephone although its frequency utilization efficiency is low andtherefore cost of call may increase. In this manner, the presentinvention system can suit for a wide variety of usage.

As is explained above, the transmission system having a distributionlike d=A+B of FIG. 118, whose capacity is locally altered, isaccomplished. Therefore, an overall frequency utilization efficiencywill be much effectively improved if layout of base stations isdetermined to fit for the actual traffic amount denoted by TF.Especially, effect of the present invention will be large in a microcell system, whose receiving cells are smaller and therefore numeroussub base stations are required. Because a large number of sub basestations can be easily installed at the place having a large trafficamount.

Next, data assignment of each time slot will be explained referring toFIG. 119, wherein FIG. 119(a) shows a conventional time slot and FIG.119(b) shows a time slot according to the eighth embodiment. Theconventional system performs a down, i.e. from a base station to aterminal station, transmission as shown in FIG. 119(a), in which a syncsignal S is transmitted by a time slot 780 a and transmission signals torespective terminal stations of A, B, C channels by time slots 780 b,780 c, 780 d respectively at a frequency A. On the other hand, an up,i.e. from the mobile station to the base station, transmission isperformed in such a manner that a sync signal S, and transmissionsignals of a, b, c channels are transmitted by time slots 781 a, 781 b,781 c, 781 d at a frequency B.

The present invention, which is characterized by a multi-level, e.g. 64SRQAM, signal transmission system, allows to have three-level dataconsisting of D₁, D₂, D₃ of 2 bit/Hz as shown in FIG. 119(b). As both ofA₁ and A₂ data are transmitted by 16 SRQAM, their time slots have twotimes data rate as shown by slots 782 b, 782 c and 783 b, 783 c. Itmeans the same quality sound can be transmitted by a half time.Accordingly, a time width of respective time slots 782 b, 782 c becomesa half. In this manner, two times transmission capacity can be acquiredat the two-level region 776 c shown in FIG. 118, i.e. at the vicinity ofthe base station.

In the same way, time slots 782 g, 783 g carry out thetransmission/reception of E1 data by use of a 64 SRQAM signal. As thetransmission capacity is three times, one time slot can be used forthree channels of E₁, E₂, E₃. This would be used for an area furtherclose to the base station. Thus, up to three times communicationcapacity can be obtained at the same frequency band. An actualtransmission efficiency, however, would be reduced to 90%. It isdesirable for enhancing the effect of the present invention to coincidethe transmission amount distribution according to the present inventionwith the regional distribution of the actual traffic amount as perfectas possible.

In fact, an actual urban area consists of a crowded building districtand a greenbelt zone surrounding this building area. Even an actualsuburb area consists of a residential district and fields or a forestsurrounding this residential district. These urban and suburb areasresemble the distribution of the TF diagram. Thus, the application ofthe present invention will be effective.

FIG. 120 is a diagram showing time slots by the TDMA method, whereinFIG. 120(a) shows a conventional method and FIG. 120(b) shows thepresent invention. The conventional method uses time slots 786 a, 786 bfor transmission to portable phones of A, B channels at the samefrequency and time slots 787 a, 787 b for transmission from the same, asshown in FIG. 120(a).

On the contrary, 16 SRQAM mode of the present invention uses a time slot788 a for reception of A₁ channel and a time slot 788 c for transmissionto A₁ channel as shown in FIG. 120(b). A width of the time slot becomesapproximately ½. In case of 64 SRQAM mode, a time slot 788 i is used forreception of D₁ channel and a time slot 7881 is used for transmission toD₁ channel. A width of the time slot becomes approximately ⅓.

In order to save electric power, a transmission of E₁ channel isexecuted by use of a normal 4 SRQAM time slot 788 r while reception ofE₁ channel is executed by use of a 16 SRQAM time slot 788 p being a ½time slot. Transmission power is surely suppressed, althoughcommunication cost may increase due to a long occupation time. This willbe effective for a small and light portable telephone equipped with asmall battery or when the battery is almost worn out.

As is described in the foregoing description, the present inventionmakes it possible to determine the distribution of transmission capacityso as to coincide with an actual traffic distribution, therebyincreasing substantial transmission capacity. Furthermore, the presentinvention allows base stations or terminal stations to freely select oneamong two or three transmission capacities. If the frequency utilizationefficiency is lowered, power consumption will be decreased. If thefrequency utilization efficiency is selected higher, communication costwill be saved. Moreover, adoption of a 4 SRQAM having smaller capacitywill simplify the circuitry and reduce the size and cost of thetelephone. As explained in the previous embodiments, one characteristicsof the present invention is that compatibility is maintained among allof associated stations. In this manner, the present invention not onlyincreases transmission capacity but allows to provide customers a widevariety of series from a super mini telephone to a high performancetelephone.

Embodiment 9

Hereinafter, a ninth embodiment of the present invention will bedescribed referring to the drawings. The ninth embodiment employs thisinvention in an OFDM transmission system. FIG. 123 is a block diagram ofan OFDM transmitter/receiver, and FIG. 124 is a diagram showing aprinciple of an OFDM action. An OFDM is one of FDM and has a betterefficiency in frequency utilization as compared with a general FDM,because an OFDM sets adjacent two carriers to be quadrate with eachother. Furthermore, OFDM can bear multipath obstruction such as ghostand, therefore, may be applied in the future to the digital musicbroadcasting or digital TV broadcasting.

As shown in the principle diagram of FIG. 124, OFDM converts an inputsignal by a serial to parallel converter 791 into a data being disposedon a frequency axis 793 at intervals of 1/ts, so as to producesubchannels 794 a˜794 e. This signal is inversely FFT converted by amodulator 4 having an inverse FFT 40 into a signal on a time axis 799 toproduce a transmission signal 795. This inverse FFT signal istransmitted during an effective symbol period 796 of the time period ts.A guard interval 797 having an amount tg is provided between symbolperiods.

A transmitting/receiving action of HDTV signal in accordance with thisninth embodiment will be explained referring to the block diagram ofFIG. 123, which shows a hybrid OFDM-CCDM system. An inputted HDTV signalis separated by a video encoder 401 into three-level, a low frequencyband D₁₋₁, a medium-low frequency band D₁₋₂, and a high-medium-lowfrequency band D₂, video signals, and fed into an input section.

In a first data stream input 743, D₁₋₁ signal is ECC encoded with highcode gain and D₁₋₂ signal is ECC coded with normal code gain. A TDM 743performs time division multiplexing of D₁₋₁ and D₁₋₂ signals to producea D₁ signal, which is then fed to a D₁ serial to parallel converter 791d in a modulator 852 a. D₁ signal consists of n pieces of parallel data,which are inputted into first inputs of n pieces of C-CDM modulator 4 a,4 b, - - - respectively.

On the other hand, the high frequency band signal D2 is fed into asecond data stream input 744 of the input section 742, in which D₂signal is ECC (Error Correction Code) encoded in an ECC 744 a and thenTrellis encoded in a Trellis encoder 744 b. Thereafter, the D₂ signal issupplied to a D₂ serial to parallel converter 791 b of the modulator 852a and converted into n pieces of parallel data, which are inputted intosecond inputs of the n pieces of C-CDM modulator 4 a, 4 b, - - -respectively.

The C-CDM modulators 4 a, 4 b, 4 c - - - respectively produces 16 SRQAMsignal on the basis of D₁ data of the first data stream input and D₂data of the second data stream input. These n pieces of C-CDM modulatorrespectively has a carrier different from each other. As shown in FIG.124, carriers 794 a, 794 b, 794 c, - - - are arrayed on the frequencyaxis 793 so that adjacent two carriers are 90°-out-of-phase with eachother. Thus C-CDM modulated n pieces of modulated signal are fed intothe inverse FFT circuit 40 and mapped from the frequency axis dimension793 to the time axis dimension 790. Thus, time signals 796 a, 796b - - - , having an effective symbol length ts, are produced. There isprovided a guard interval zone 797 a of Tg seconds between the effectivesymbol time zones 796 a and 796 b, in order to reduce multipathobstruction. FIG. 129 is a graph showing a relationship between timeaxis and signal level. The guard time Tg of the guard interval band 797a is determined by taking account of multipath affection and usage ofsignal. By setting the guard time Tg longer than the multipath affectedtime, e.g. TV ghost, modulated signals from the inverse FFT circuit 40are converted by a parallel to serial converter 4 e into one signal and,then, transmitted from a transmitting circuit 5 as an RF signal.

Next, an action of a receiver 43 will be described. A received signal,shown as time-base symbol signal 796 e of FIG. 124, is fed into an inputsection 24 of FIG. 123. Then, the received signal is converted into adigital signal in a demodulator 852 b and further changed into Fouriercoefficients in a FFT 40 a. Thus, the signal is mapped from the timeaxis 799 to the frequency axis 793 a as shown in FIG. 124. That is, thetime-base symbol signal is converted into frequency-base carriers 794 a,794 b, - - - . As these carriers are in quadrature relationship witheach other, it is possible to separate respective modulated signals.FIG. 125(b) shows thus demodulated 16 SRQAM signal, which is then fed torespective C-CDM demodulators 45 a, 45 b, - - - of a C-CDM demodulator45, in which demodulated 16 SRQAM signal is demodulated into multi-levelsub signals D₁, D₂. These sub signals D₁ and D₂ are further demodulatedby a D₁ parallel to serial converter 852 a and a D₂ parallel to serialconverter 852 b into original D₁ and D₂ signals.

Since the signal transmission system is of C-CDM multi-level shown in125(b), both D₁ and D₂ signals will be demodulated under betterreceiving condition but only D₁ signal will be demodulated under worse,e.g. low C/N rate, receiving condition. Demodulated D₁ signal isdemodulated in an output section 757. As D₁₋₁ signal has higher ECC codegain as compared with the D₁₋₂ signal, an error signal of the D₁₋₁signal is reproduced even under worse receiving condition.

The D₁₋₁ signal is converted by a 1-1 video decoder 402 c into a lowfrequency band signal and outputted as an LDTV, and the D₁₋₂ signal isconverted by a 1-2 video decoder 402 d into a medium frequency bandsignal and outputted as EDTV.

The D₂ signal is Trellis decoded by a Trellis decoder 759 b andconverted by a second video decoder 402 b into a high frequency bandsignal and outputted as an HDTV signal. Namely, an LDTV signal isoutputted in case of the low frequency band signal only. An EDTV signalof a wide NTSC grade is outputted if the medium frequency band signal isadded to the low frequency band signal, and an HDTV signal is producedby adding low, medium, and high frequency band signals. As well as theprevious embodiment, a TV signal having a picture quality depending on areceiving C/N rate can be received. Thus, the ninth embodiment realizesa novel multi-level signal transmission system by combining an OFDM anda C-CDM, which was not obtained by the OFDM alone.

An OFDM is certainly strong against multipath such as TV ghost becausethe guard time Tg can absorb an interference signal of multipath.Accordingly, the OFDM is applicable to the digital TV broadcasting forautomotive vehicle TV receivers. Meanwhile, no OFDM signal is receivedwhen the C/N rate is less than a predetermined value because its signaltransmission pattern is non of a multi-level type.

However the present invention can solve this disadvantage by combiningthe OFDM with the C-CDM, thus realizing a graditional degradationdepending on the C/N rate in a video signal reception without beingdisturbed by multipath.

When a TV signal is received in a compartment of vehicle, not only thereception is disturbed by multipath but the C/N rate is deteriorated.Therefore, the broadcast service area of a TV broadcast station will notbe expanded as expected if the countermeasure is only for multipath.

On the other hand, a reception of TV signal of at least LDTV grade willbe ensured by the combination with the multi-level transmission C-CDMeven if the C/N rate is fairly deteriorated. As a picture plane size ofan automotive vehicle TV is normally less than 10 inches, a TV signal ofan LDTV grade will provide a satisfactory picture quality. Thus, theLDTV grade service area of automotive vehicle TV will largely expanded.If an OFDM is used in an entire frequency band of HDTV signal, presentsemiconductor technologies cannot prevent circuitry scale fromincreasing so far.

Now, an OFDM method of transmitting only D₁₋₁ of low frequency band TVsignal will be explained below. As shown in a block diagram in FIG. 138,a medium frequency band component D₁₋₂ and a high frequency bandcomponent D₂ of an HDTV signal are multiplexed in C-CDM modulator 4 a,and then transmitted at a frequency band A through an FDM 40 d.

On the other hand, a signal received by a receiver 43 is first of allfrequency separated by an FDM 40 e and, then, demodulated by a C-CDMdemodulator 4 b of the present invention. Thereafter, thus C-CDMdemodulated signal is reproduced into medium and high frequencycomponents of HDTV in the same way as in FIG. 123. An operation of avideo decoder 402 is identical to that of embodiments 1, 2, and 3 andwill no more be explained.

Meanwhile, the D₁₋₁ signal, a low frequency band signal of MPEG 1 gradeof HDTV, is converted by a serial to parallel converter 791 into aparallel signal and fed to an OFDM modulator 852 c, which executes QPSKor 16 QAM modulation. Subsequently, the D₁₋₁ signal is converted by aninverse FFT 40 into a time-base signal and transmitted at a frequencyband B through a FDM 40 d.

On the other hand, a signal received by the receiver 43 is frequencyseparated in the FDM 40 e and, then, converted into a number offrequency-base signals in an FFT 40 a of an OFDM modulator 852 d.Thereafter, frequency-base signals are demodulated in respectivedemodulators 4 a, 4 b, - - - and are fed into a parallel to serialconverter 882 a, wherein a D₁₋₁ signal is demodulated. Thus, a D₁₋₁signal of LDTV grad is outputted from the receiver 43.

In this manner, only an LDTV signal is OFDM modulated in the multi-levelsignal transmission. The method of FIG. 138 makes it possible to providea complicated OFDM circuit only for an LDTV signal. A bit rate of LDTVsignal is 1/20 of that of an HDTV. Therefore, the circuit scale of theOFDM will be reduced to 1/20, which results in an outstanding reductionof overall circuit scale.

An OFDM signal transmission system is strong against multipath and willsoon be applied to a moving station, such as a portable TV, anautomotive vehicle TV, or a digital music broadcast receiver, which isexposed under strong and variable multipath obstruction. For such usagesa small picture size of less than 10 inches, 4 to 8 inches, is themainstream. It will be thus guessed that the OFDM modulation of a highresolution TV signal such as HDTV or EDTV will bring less effect. Inother words, the reception of a TV signal of LDTV grade would besufficient for an automotive vehicle TV.

On the contrary, multipath is constant at a fixed station such as a homeTV. Therefore, a countermeasure against multipath is relatively easy.Less effect will be brought to such a fixed station by OFDM unless it isin a ghost area. Using OFDM for medium and high frequency bandcomponents of HDTV is not advantageous in view of present circuit scaleof OFDM which is still large.

Accordingly, the method of the present invention, in which OFDM is usedonly for a low frequency band TV signal as shown in FIG. 138, can widelyreduce the circuit scale of the OFDM to less than 1/10 without losinginherent OFDM effect capable of largely reducing multiple obstruction ofLDTV when received at a mobile station such as an automotive vehicle.

Although the OFDM modulation of FIG. 138 is performed only for D₁₋₁signal, it is also possible to modulate both D₁₋₁ and D₁₋₁ by OFDM. Insuch a case, a C-CDM two-level signal transmission is used fortransmission of D₁₋₁ and D₁₋₂. Thus, a multi-level broadcasting beingstrong against multipath will be realized for a vehicle such as anautomotive vehicle. Even in a vehicle, the gradational graduation willbe realized in such a manner that LDTV and SDTV signals are receivedwith picture qualities depending on receiving signal level or antennasensitivity.

The multi-level signal transmission according to the present inventionis feasible in this manner and produces various effects as previouslydescribed. Furthermore, if the multi-level signal transmission of thepresent invention is incorporated with an OFDM, it will become possibleto provide a system strong against multipath and to alter datatransmission grade in accordance with receivable signal level change.

FIG. 126(a) shows another method of realizing the multi-level signaltransmission system, wherein the subchannels 794 a-794 c of the OFDM areassigned to a first layer 801 a and the subchannels 794 d-794 f areassigned to a second layer 801 b. There is provided a frequency guardzone 802 a of f_(g) between these two, first and second, layers. FIG.126(b) shows an electric power difference 802 b of Pg which is providedto differentiate the transmission power of the first and second layers801 a and 801 b.

Utilization of this differentiation makes it possible to increaseelectric power of the first layer 801 a in the range not obstructing theanalogue TV broadcast service as shown in FIG. 108(d) previouslydescribed. In this case, a threshold value of the C/N ratio capable ofreceiving the first layer 801 a becomes lower than that for the secondlayer 801 b as shown in FIG. 108(e). Accordingly, the first layer 801 acan be received even in a low signal-level area or in a large-noisearea. Thus, a two-layer signal transmission is realized as shown in FIG.147. This is referred to as Power-Weighted-OFDM system (i.e. PW-OFDM) inthis specification. If this PW-OFDM system is combined with the C-CDMsystem previously explained, three layers will be realized as shown inFIG. 108(e) and, accordingly, the signal receivable area will becorrespondingly expanded.

FIG. 144 shows a specific circuit, wherein the first layer data passingthrough the first data stream circuit 791 a is modulated into thecarriers f₁-f₃ by the modulators 4 a-4 c having large amplitude and,then, are OFDM modulated in the inverse FFT 40. On the contrary, thesecond layer data passing through the second data stream circuit 791 bis modulated into the carriers f₆-f₈ by the modulators 4 d-4 f havingordinary amplitude and, then, are OFDM modulated in the inverse FFT 40.Then, these OFDM modulated signals are transmitted from the transmitcircuit 5.

A signal received by the receiver 43 is separated into several signalshaving carriers of f₁-f_(n) through the FFT 40 a. The carriers f₁-f₃ aredemodulated by the demodulators 45 a-45 c to reproduce the first datastream D₁, i.e. the first layer 801 a. On the other hand, the carriersf₆-f₈ are demodulated by the demodulators 45 d-45 f to reproduce thesecond data stream D₂, i.e. the second layer 801 b.

The first layer 801 a has so large electric power that it can bereceived even in a weak-signal area. In this manner, the PW-OFDM systemrealizes the two-layer multi-level signal transmission. If this PW-OFDMis combined with the C-CDM, it will become possible to provide 3-4layers. As the circuit of FIG. 144 is identical with the circuit of FIG.123 in the remaining operations and, therefore, will no more beexplained.

Next, a method of realizing a multi-level signal transmission inTime-Weighted-OFDM (i.e. TW-OFDM) in accordance with the presentinvention will be explained. Although the OFDM system is accompaniedwith the guard time zone t_(g) as previously described, adverseaffection of ghost will be eliminated if the delay time t_(H) of theghost, i.e. multipath, signal satisfies the requirement of t_(M)<t_(g).The delay time t_(M) will be relatively small, for example in the rangeof several μs, in a fixed station such as a TV receiver used for homeuse. Furthermore, as its value is constant, cancellation of ghost willbe relatively easily done. On the contrary, reflected wave will increasein case of a mobile station such as a vehicle TV receiver. Therefore,the delay time t_(M) becomes relatively large, for example in the rangeof several tens μs. Furthermore, the magnitude of t_(M) varies inresponse to the running movement of the vehicle. Thus, cancellation ofghost tends to be difficult. Hence, the multi-level signal transmissionis key or essential for such a mobile station TV receiver in order toeliminate adverse affection of multipath.

The multi-level signal transmission in accordance with the presentinvention will be explained below. A symbol contained in the subchannellayer A can be intensified against the ghost by setting a guard timet_(ga) of the layer A to be larger than a guard time t_(gb) of the layerB as shown in FIG. 146. In this manner, the multi-layer signaltransmission can be realized against multipath by use of weighting ofguard time. This system is referred to as Guard-Time-Weighted-OFDM (i.e.QTW-OFDM).

If the symbol number of the symbol time Ts is not different in the layerA and in the layer B, a symbol time t_(sa) of the layer A is set to belarger than a symbol time t_(sb) of the layer B. With thisdifferentiation, a carrier width Δfa of the carrier A becomes largerthan a carrier width Δfb of the carrier B. (Δfa>Δfb) Therefore, theerror rate becomes lower in the demodulation of the symbol of the layerA compared with the demodulation of the symbol of the layer B. Thus, thedifferentiation of the layers A and B in the weighting of the symboltime Ts can realize a two-layer signal transmission against multipath.This system is referred to as Carrier-Spacing-Weighted-OFDM (i.e.CSW-OFDM).

By realizing the two-layer signal transmission based on the GTW-OFDM,wherein a low-resolution TV signal is transmitted by the layer A and ahigh-frequency component is transmitted by the layer B, the vehicle TVreceiver can stably receive the low-resolution TV signal regardless oftough ghost. Furthermore, the multi-level signal transmission withrespect to the C/N ratio can be realized by differentiating the symboltime t_(s) based on the CSW-OFDM between the layers A and B. If thisCSW-OFDM is combined with the GTW-OFDM, the signal reception in thevehicle TV receiver can be further stabilized. High resolution is notnormally required to the vehicle TV or the portable TV.

As the time ratio of the symbol time including a low-resolution TVsignal is small, an overall transmission efficiency will not decrease somuch even if the guard time is enlarged. Accordingly, using the GTW-OFDMof the present invention for suppressing multipath by laying emphasis onthe low-resolution TV signal will realize the multi-layer type TVbroadcast service wherein the mobile station such as the portable orvehicle TV receiver can be compatible with the stationary station suchas the home TV without substantially lowering the transmissionefficiency. If combined with the CSW-OFDM or the C-CDM as describedpreviously, the multi-layer to the C/N ratio can be also realized. Thus,the signal reception in the mobile station will be further stabilized.

An affection of the multipath will be explained in more detail. In caseof multipaths 810 a, 810 b, 810 c, and 810 d having shorter delay timeas shown in FIG. 145(a), the signals of both the first and second layerscan be received and therefore the HDTV signal can be demodulated. On thecontrary, in case of multipaths 811 a, 811 b, 811 c, and 811 d havinglonger delay time as shown in FIG. 145(b), the B signal of the secondlayer cannot be received since its guard time t_(gb) is not sufficientlylong. However, the A signal of the first layer can be received withoutbeing bothered by the multipath since its guard time t_(ga) issufficiently long. As described above, the B signal includes thehigh-frequency component of TV signal. The A signal includes thelow-frequency component of TV signal. Accordingly, the vehicle TV canreproduce the LDTV signal. Furthermore, as the symbol time Tsa is setlarger than symbol time Tsb, the first layer is strong againstdeterioration of C/N ratio.

Such a discrimination of the guard time and the symbol time is effectiveto realize two-dimensional multi-layer signal transmission of the OFDMin a simple manner. If the discrimination of guard time is combined withthe C-CDM in the circuit shown in FIG. 123, the multi-layer signaltransmission effective against both multipath and deterioration of C/Nratio will be realized.

Next, a specific example will be described below.

The smaller the D/U ratio of the receiving signal becomes, the largerthe multipath delay time T_(M) becomes. Because, the reflected waveincreases compared with the direct wave. For example, as shown in FIG.148, if the D/U ratio is smaller than 30 dB, the delay time T_(M)exceeds 30 μs because of increase of the reflected wave. Therefore, ascan be understood from FIG. 148, it will become possible to receive thesignal even in the worst condition if the Tg is set to be larger than 50μs.

Accordingly, as shown in detail in FIGS. 149(a) and 149(b), three groupsof first 801 a, second 801 b, and third 801 c layers are assigned in a 2ms period of 1 sec TV signal. The guard times 797 a, 797 b, and 797 c,i.e. Tga, Tgb, and Tgc, of these three groups are weighted to be, forexample, 50 μs, 5 μs, and 1 μs, respectively, as shown in FIG. 149(c).Thus, three-layer signal transmission effective to the multipath will berealized as shown in FIG. 150, wherein three layers 801 a, 801 b, and801 c are provided.

If the GTW-OFDM is applied to all the picture quality, it is doubtlessthat, the transmission efficiency will decrease. However, if theGTW-OFDM is only applied to the LDTV signal including less informationfor the purpose of suppression of multipath, it is expected that anoverall transmission efficiency will not be worsened so much.Especially, as the first layer 801 a has a long guard time Tg of 50 μslarger than 30 μs, it will be received even by the vehicle TV receiver.The circuit shown in FIG. 127 will be suitable for this purpose.Especially, the requirement to the quality of vehicle TV is LDTV grade.Therefore, its transmission capacity will be approximately 1 Mbps ofMPEG 1 class. If the symbol time 796 a, i.e. Tsa, is set to be 200 μswith respect to the 2 ms period as shown in FIG. 149, the transmissioncapacity becomes 2 Mbps. Even if the symbol rate is lowered less thanhalf, an approximately 1 Mbps capacity can be kept. Therefore, it ispossible to ensure picture quality of LDTV grade. Although thetransmission efficiency is slightly decreased, the error rate can beeffectively lowered by the CSW-OFDM in accordance with the presentinvention. If the C-CDM of the present invention is combined with theGTW-OFDM, deterioration of the transmission efficiency will be able tobe effectively prevented. In FIG. 149, the symbol times 796 a, 796 b,and 796 c of the same symbol number are differentiated to be 200 μs, 150μs, and 100 μs, respectively. Accordingly, the error rate becomes highin the order of the first, second, and third layers so as to realize themulti-layer signal transmission.

At the same time, the multi-layer signal transmission effective to C/Nratio can be realized. By combining the CSW-OFDM and the CSW-OFDM, atwo-dimensional multi-layer signal transmission is realized with respectto the multipath and the C/N ratio as shown in FIG. 151. As describedpreviously, it is possible to combine the CSW-OFDM and the C-CDM of thepresent invention for preventing the overall transmission efficiencyfrom being lowered. In the first, 1-2, and 1-3 layers 801 a, 851 a, and851 az, the LDTV grade signal can be stably received by, for example,the vehicle TV receiver subjected to the large multipath T_(M) and lowC/N ratio. In the second and 2-3 layers 801 b and 851 b, thestandard-resolution SDTV grade signal can be received by the fixed orstationary station located, for example, in the fringe of the servicearea which is generally subjected to the lower C/N ratio and ghost. Inthe third layer 801 c which occupies more than half of the service area,the HDTV grade signal can be received since the C/N ratio is high andthe ghost is less because of large direct wave. In this manner, atwo-dimensional multi-layer broadcast service effective to both the C/Nratio and the multipath can be realized by the combination of theGTW-OFDM and the C-CDM or the combination of the GTW-OFDM and theCSW-C-CDM in accordance with the present invention. Thus, the presentinvention realizes a two-dimensional, matrix type, multi-layer signaltransmission system effective to both the C/N ratio and the multipath,which has not ever been realized by the prior art technologies.

The multi-level signal transmission method of the present invention isintended to increase the utilization of frequencies but may be suitedfor not all the transmission systems since causing some type receiversto be declined in the energy utilization. It is a good idea for use witha satellite communications system for selected subscribers to employmost advanced transmitters and receivers designed for best utilizationof applicable frequencies and energy. Such a specific purpose signaltransmission system will not be bound by the present invention.

The present invention will be advantageous for use with a satellite orterrestrial broadcast service which is essential to run in the samestandards for as long as 50 years. During the service period, thebroadcast standards must not be altered but improvements will beprovided time to time corresponding to up-to-date technologicalachievements. Particularly, the energy for signal transmission willsurely be increased on any satellite. Each TV station should provide acompatible service for guaranteeing TV program signal reception to anytype receivers ranging from today's common ones to future advanced ones.The signal transmission system of the present invention can provide acompatible broadcast service of both the existing NTSC and HDTV systemsand also, ensure a future extension to match mass data transmission.

The present invention concerns much on the frequency utilization thanthe energy utilization. The signal receiving sensitivity of eachreceiver is arranged different depending on a signal state level to bereceived so that the transmitting power of a transmitter needs not beincreased largely. Hence, existing satellites which offer a small energyfor reception and transmission of a signal can best be used with thesystem of the present invention. The system is also arranged forperforming the same standards corresponding to an increase in thetransmission energy in the future and offering the compatibility betweenold and new type receivers. In addition, the present invention will bemore advantageous for use with the satellite broadcast standards.

The multi-level signal transmission method of the present invention ismore preferably employed for terrestrial TV broadcast service in whichthe energy utilization is not crucial, as compared with satellitebroadcast service. The results are such that the signal attenuatingregions in a service area which are attributed to a conventional digitalHDTV broadcast system are considerably reduced in extension and also,the compatibility of an HDTV receiver or display with the existing NTSCsystem is obtained. Furthermore, the service area is substantiallyincreased so that program suppliers and sponsors can appreciate moreviewers. Although the embodiments of the present invention refer to 16and 32 QAM procedures, other modulation techniques including 64, 128,and 256 QAM will be employed with equal success. Also, multiple PSK,ASK, and FSK techniques will be applicable as described with theembodiments.

A combination of the TDM with the SRQAM of the present invention hasbeen described in the above. However, the SRQAM of the present inventioncan be combined also with any of the FDM, CDMA and frequency dispersalcommunications systems.

1-8. (canceled)
 9. A signal transmission apparatus comprising: amodulator operable to modulate a first data stream according to anm-level PSK or an m-level QAM and modulate a second data streamaccording to an n-level PSK or an n-level QAM to produce modulatedsignals, wherein the first data stream includes information representingthe value of n, and n is an integer and equal to or greater than m; aconverter operable to convert the modulated signals into a convertedsignal having an effective symbol part and a guard interval, accordingto Orthogonal Frequency Division Multiplexing, and a transmitteroperable to transmit the converted signal, wherein the guard interval isselected from a plurality of predetermined time periods.
 10. A signalreceiving apparatus comprising: a receiver operable to receive an inputsignal to produce a received signal; wherein the input signal having aneffective symbol part and a guard interval, the input signal beingconverted according to Orthogonal Frequency Division Multiplexing, theinput signal having information of a first data stream and a second datastream, the first data stream being modulated an m-level PSK or anm-level QAM, the second data stream being modulated an n-level PSK or ann-level QAM, the first data stream having information representing thevalue of n, n being an integer and equal to or greater than m, and theguard interval being selected from a plurality of predetermined timeperiods; a converter operable to convert the received signal into aconverted first data stream and a converted second data stream,according to Orthogonal Frequency Division Multiplexing; and ademodulator operable to demodulate the converted first data stream toproduce the first data stream and demodulate the converted second datastream to produce the second data stream, wherein the second data streamis produced according to the value of n.
 11. A signal transmissionsystem comprising: a signal transmission apparatus comprising: amodulator operable to modulate a first data stream according to anm-level PSK or an m-level QAM and modulate a second data streamaccording to an n-level PSK or an n-level QAM to produce modulatedsignals, wherein the first data stream includes information representingthe value of n, and n is an integer and equal to or greater than m; afrequency-time converter operable to convert the modulated signals intoa converted signal having an effective symbol part and a guard interval,according to Orthogonal Frequency Division Multiplexing, and wherein theguard interval is selected from a plurality of predetermined timeperiods; a transmitter operable to transmit the converted signal; asignal receiving apparatus comprising: a time-frequency converteroperable to convert the converted signal into a converted first datastream and a converted second data stream, according to OrthogonalFrequency Division Multiplexing; and a demodulator operable todemodulate the converted first data stream to produce the first datastream and demodulate the converted second data stream to produce thesecond data stream; wherein the second data stream is produced accordingto the value of n.
 12. A signal transmission method comprising:modulating a first data stream according to an m-level PSK or an m-levelQAM and modulating a second data stream according to an n-level PSK oran n-level QAM to produce modulated signals, wherein the first datastream includes information representing the value of n, and n is aninteger and equal to or greater than m; converting the modulated signalsinto a converted signal having an effective symbol part and a guardinterval, according to Orthogonal Frequency Division Multiplexing; andtransmitting the converted signal; wherein the guard interval isselected from a plurality of predetermined time periods.
 13. A signalreceiving method comprising: receiving an input signal to produce areceived signal; wherein the input signal having an effective symbolpart and a guard interval, the input signal being converted according toOrthogonal Frequency Division Multiplexing, the input signal havinginformation of a first data stream and a second data stream, the firstdata stream being modulated an m-level PSK or an m-level QAM, the seconddata stream being modulated an n-level PSK or an n-level QAM, the firstdata stream having information representing the value of n, n being aninteger and equal to or greater than m, and the guard interval beingselected from a plurality of predetermined time periods; converting thereceived signal into a converted first data stream and a convertedsecond data stream, according to Orthogonal Frequency DivisionMultiplexing; and demodulating the converted first data stream toproduce the first data stream and demodulating the converted second datastream to produce the second data stream; wherein the second data streamis produced according to the value of n.
 14. A signal transmission andreceiving method comprising a signal transmission method and a signalreceiving method, a signal transmission method comprising: modulating afirst data stream according to an m-level PSK or an m-level QAM andmodulating a second data stream according to an n-level PSK or ann-level QAM to produce modulated signals, wherein the first data streamincludes information representing the value of n, and n is an integerand equal to or greater than m; converting the modulated signals into aconverted signal having an effective symbol part and a guard interval,according to Orthogonal Frequency Division Multiplexing; and wherein theguard interval is selected from a plurality of predetermined timeperiods; transmitting the converted signal; a signal receiving methodcomprising: converting the converted signal into a converted first datastream and a converted second data stream, according to OrthogonalFrequency Division Multiplexing; and demodulating the converted firstdata stream to produce the first data stream and demodulating theconverted second data stream to produce the second data stream; whereinthe second data stream is produced according to the value of n.